Digital broadcasting system and method of processing data

ABSTRACT

A digital broadcast system and method of processing data are disclosed. A channel equalizer includes a frequency domain converter receiving a known data sequence, when the known data sequence is periodically inserted and transmitted in general data, and converting the received data to frequency domain data, a CIR estimator using the data being received during a known data section and known data generated by a receiving system, so as to estimate a CIR, a CIR calculator interpolating or extrapolating the CIR estimated by the CIR estimator in accordance with characteristics of the general data being received, a coefficient calculator converting the CIR being outputted from the CIR calculator to a frequency domain CIR and calculating and outputting an equalization coefficient, and a distortion compensator multiplying the equalization coefficient calculated by the coefficient calculator with the data converted to frequency domain data by the frequency domain converter, thereby compensating channel distortion.

This application claims the benefit of the Korean Patent Application No.10-2007-0031960, filed on Mar. 30, 2007, which is hereby incorporated byreference as if fully set forth herein. Also, this application claimsthe benefit of U.S. Provisional Application No. 60/909,575, filed onApr. 2, 2007, which is hereby incorporated by reference.

BACKGROUND OF THE INVENTION

1. Field of the Invention

The present invention relates to a digital broadcasting system andmethod of processing data.

2. Discussion of the Related Art

The Vestigial Sideband (VSB) transmission mode, which is adopted as thestandard for digital broadcasting in North America and the Republic ofKorea, is a system using a single carrier method. Therefore, thereceiving performance of the digital broadcast receiving system may bedeteriorated in a poor channel environment. Particularly, sinceresistance to changes in channels and noise is more highly required whenusing portable and/or mobile broadcast receivers, the receivingperformance may be even more deteriorated when transmitting mobileservice data by the VSB transmission mode.

SUMMARY OF THE INVENTION

Accordingly, the present invention is directed to a digital broadcastingsystem and a method of processing data that substantially obviate one ormore problems due to limitations and disadvantages of the related art.

An object of the present invention is to provide a digital broadcastingsystem and a method of processing data that are highly resistant tochannel changes and noise.

Another object of the present invention is to provide a digitalbroadcasting system and a method of processing data that can enhance thereceiving performance of a digital broadcast receiving system byperforming additional encoding on mobile service data and bytransmitting the processed data to the receiving system.

A further object of the present invention is to provide a digitalbroadcasting system and a method of processing data that can alsoenhance the receiving performance of a digital broadcast receivingsystem by inserting known data already known in accordance with apre-agreement between the receiving system and the transmitting systemin a predetermined area within a data area.

Additional advantages, objects, and features of the invention will beset forth in part in the description which follows and in part willbecome apparent to those having ordinary skill in the art uponexamination of the following or may be learned from practice of theinvention. The objectives and other advantages of the invention may berealized and attained by the structure particularly pointed out in thewritten description and claims hereof as well as the appended drawings.

To achieve these objects and other advantages and in accordance with thepurpose of the invention, as embodied and broadly described herein, adigital broadcast transmitting system includes a service multiplexer anda transmitter. The service multiplexer multiplexes mobile service dataand main service data at pre-determined data rates and, then, transmitsthe multiplexed service data to the transmitter. The transmitterperforms additional encoding on the mobile service data transmitted fromthe service multiplexer and, also, groups a plurality of mobile servicedata packets having encoding performed thereon so as to configure a datagroup.

Herein, the transmitter may multiplex a mobile service data packetincluding the mobile service data and a main service data packetincluding the main service data in packet units and may transmit themultiplexed data packets to a digital broadcast receiving system.Herein, the transmitter may multiplex the data group and the mainservice data packet in a burst structure, wherein the burst section maybe divided in a burst-on section including the data group, and aburst-off section that does not include the data group. The data groupmay be divided into a plurality of regions based upon a degree ofinterference of the main service data. A long known data sequence may beperiodically inserted in the region having no interference with the mainservice data.

In another aspect of the present invention, a digital broadcastreceiving system may use the known data sequence for demodulating andchannel equalizing processes. When receiving only the mobile servicedata, the digital broadcast receiving system turns power on only duringthe burst-on section so as to process the mobile service data.

In another aspect of the present invention, a channel equalizer of adigital broadcast receiving system includes a frequency domainconverter, a CIR estimator, a CIR calculator, a coefficient calculator,and a distortion compensator. Herein, the frequency domain converterreceives a known data sequence, when the known data sequence isperiodically inserted and transmitted in general data, and converts thereceived data to frequency domain data. The CIR estimator uses the databeing received during a known data section and known data generated by areceiving system, so as to estimate a channel impulse response (CIR).The CIR calculator interpolates or extrapolates the CIR estimated by theCIR estimator in accordance with characteristics of the general databeing received. The coefficient calculator converts the CIR beingoutputted from the CIR calculator to a frequency domain CIR andcalculates and outputs an equalization coefficient. And, the distortioncompensator multiplies the equalization coefficient calculated by thecoefficient calculator with the data converted to frequency domain databy the frequency domain converter, thereby compensating channeldistortion.

It is to be understood that both the foregoing general description andthe following detailed description of the present invention areexemplary and explanatory and are intended to provide furtherexplanation of the invention as claimed.

BRIEF DESCRIPTION OF THE DRAWINGS

The accompanying drawings, which are included to provide a furtherunderstanding of the invention and are incorporated in and constitute apart of this application, illustrate embodiment(s) of the invention andtogether with the description serve to explain the principle of theinvention. In the drawings:

FIG. 1 illustrates a block diagram showing a general structure of adigital broadcasting system according to an embodiment of the presentinvention;

FIG. 2 illustrates a block diagram showing an example of a servicemultiplexer of FIG. 1;

FIG. 3 illustrates a block diagram showing an example of a transmitterof FIG. 1;

FIG. 4 illustrates a block diagram showing an example of a pre-processorof FIG. 3;

FIG. 5( a) to FIG. 5( e) illustrate error correction encoding and errordetection encoding processed according to an embodiment of the presentinvention;

FIG. 6A and FIG. 6B illustrate data configuration before and after adata deinterleaver in a digital broadcast transmitting system accordingto the present invention;

FIG. 7 illustrates a process of dividing a RS frame for configuring adata group according to the present invention;

FIG. 8 illustrates exemplary operations of a packet multiplexer fortransmitting the data group according to the present invention;

FIG. 9 illustrates a block diagram showing a structure of a blockprocessor according to the present invention;

FIG. 10 illustrates a detailed block diagram of a symbol encoder shownin FIG. 9;

FIG. 11( a) to FIG. 11( c) illustrate a variable length interleavingprocess of a symbol interleaver shown in FIG. 9;

FIG. 12A and FIG. 12B illustrate block diagrams showing structures of ablock processor according to another embodiment of the presentinvention;

FIG. 13( a) to FIG. 13( c) illustrate block encoding and trellisencoding processes according to the present invention;

FIG. 14 illustrates a block diagram showing a trellis encoding moduleaccording to the present invention;

FIG. 15A and FIG. 15B a block processor and a trellis encoding moduleconnected to one another according to the present invention;

FIG. 16 illustrates a block processor according to another embodiment ofthe present invention;

FIG. 17 illustrates a block processor according to yet anotherembodiment of the present invention;

FIG. 18 illustrates an example of a group formatter inserting andtransmitting a transmission parameter;

FIG. 19 illustrates an example of a block processor inserting andtransmitting a transmission parameter;

FIG. 20 illustrates an example of a packet formatter inserting andtransmitting a transmission parameter;

FIG. 21 illustrates an example for inserting and transmitting thetransmission parameter in a field synchronization segment area;

FIG. 22 illustrates a block diagram showing a structure of a digitalbroadcast receiving system according to the present invention;

FIG. 23 illustrates an example of known data sequence being periodicallyinserted and transmitted in-between actual data by a transmittingsystem;

FIG. 24( a) to FIG. 24( c) illustrate correlation between a main signaland a ghost signal when a post-ghost exists;

FIG. 25( a) to FIG. 25( c) illustrate correlation between a main signaland a ghost signal when a pre-ghost exists;

FIG. 26 illustrates a block diagram of an example of a channel equalizeraccording to the present invention;

FIG. 27 illustrates a block diagram of another example of a channelequalizer according to the present invention;

FIG. 28 illustrates a block diagram of another example of a channelequalizer according to the present invention;

FIG. 29 illustrates a block diagram of another example of a channelequalizer according to the present invention;

FIG. 30 illustrates a block diagram of another example of a channelequalizer according to the present invention;

FIG. 31 illustrates a block diagram of another example of a channelequalizer according to the present invention;

FIG. 32 illustrates an example of an interpolation operation;

FIG. 33 illustrates an example of overlap & save processes according toan embodiment of the present invention;

FIG. 34 illustrates another example of overlap & save processesaccording to the present invention;

FIG. 35 illustrates an example of an extrapolation operation;

FIG. 36 illustrates a conceptual diagram of the extrapolation operationaccording to the present invention;

FIG. 37 illustrates a block diagram of another example of a channelequalizer according to the present invention;

FIG. 38 illustrates a block diagram of another example of a channelequalizer according to the present invention;

FIG. 39 illustrates a block diagram of another example of a channelequalizer according to the present invention;

FIG. 40 illustrates a block diagram of another example of a channelequalizer according to the present invention;

FIG. 41 illustrates a block diagram of an example of a CIR estimatoraccording to the present invention;

FIG. 42 illustrates a block diagram of another example of a channelequalizer according to the present invention;

FIG. 43 illustrates a detailed block diagram of an example of aremaining carrier phase error estimator according to the presentinvention;

FIG. 44 illustrates a block diagram of a phase error detector obtaininga remaining carrier phase error and phase noise according to the presentinvention;

FIG. 45 illustrates a phase compensator according to an embodiment ofthe present invention; and

FIG. 46( a) to FIG. 46( g), FIG. 46( a′) and FIG. 46( b′) illustrates anexample of an error correction decoding process according to the presentinvention.

DETAILED DESCRIPTION OF THE INVENTION

Reference will now be made in detail to the preferred embodiments of thepresent invention, examples of which are illustrated in the accompanyingdrawings. Wherever possible, the same reference numbers will be usedthroughout the drawings to refer to the same or like parts. In addition,although the terms used in the present invention are selected fromgenerally known and used terms, some of the terms mentioned in thedescription of the present invention have been selected by the applicantat his or her discretion, the detailed meanings of which are describedin relevant parts of the description herein. Furthermore, it is requiredthat the present invention is understood, not simply by the actual termsused but by the meaning of each term lying within.

Among the terms used in the description of the present invention, mainservice data correspond to data that can be received by a fixedreceiving system and may include audio/video (A/V) data. Morespecifically, the main service data may include A/V data of highdefinition (HD) or standard definition (SD) levels and may also includediverse data types required for data broadcasting. Also, the known datacorrespond to data pre-known in accordance with a pre-arranged agreementbetween the receiving system and the transmitting system. Additionally,in the present invention, mobile service data may include at least oneof mobile service data, pedestrian service data, and handheld servicedata, and are collectively referred to as mobile service data forsimplicity. Herein, the mobile service data not only correspond tomobile/pedestrian/handheld service data (M/P/H service data) but mayalso include any type of service data with mobile or portablecharacteristics. Therefore, the mobile service data according to thepresent invention are not limited only to the M/P/H service data.

The above-described mobile service data may correspond to data havinginformation, such as program execution files, stock information, and soon, and may also correspond to A/V data. Most particularly, the mobileservice data may correspond to A/V data having lower resolution andlower data rate as compared to the main service data. For example, if anA/V codec that is used for a conventional main service corresponds to aMPEG-2 codec, a MPEG-4 advanced video coding (AVC) or scalable videocoding (SVC) having better image compression efficiency may be used asthe NV codec for the mobile service. Furthermore, any type of data maybe transmitted as the mobile service data. For example, transportprotocol expert group (TPEG) data for broadcasting real-timetransportation information may be serviced as the main service data.

Also, a data service using the mobile service data may include weatherforecast services, traffic information services, stock informationservices, viewer participation quiz programs, real-time polls & surveys,interactive education broadcast programs, gaming services, servicesproviding information on synopsis, character, background music, andfilming sites of soap operas or series, services providing informationon past match scores and player profiles and achievements, and servicesproviding information on product information and programs classified byservice, medium, time, and theme enabling purchase orders to beprocessed. Herein, the present invention is not limited only to theservices mentioned above. In the present invention, the transmittingsystem provides backward compatibility in the main service data so as tobe received by the conventional receiving system. Herein, the mainservice data and the mobile service data are multiplexed to the samephysical channel and then transmitted.

The transmitting system according to the present invention performsadditional encoding on the mobile service data and inserts the dataalready known by the receiving system and transmitting system (i.e.,known data), thereby transmitting the processed data. Therefore, whenusing the transmitting system according to the present invention, thereceiving system may receive the mobile service data during a mobilestate and may also receive the mobile service data with stabilitydespite various distortion and noise occurring within the channel.

General Description of a Transmitting System

FIG. 1 illustrates a block diagram showing a general structure of adigital broadcast transmitting system according to an embodiment of thepresent invention. Herein, the digital broadcast transmitting includes aservice multiplexer 100 and a transmitter 200. Herein, the servicemultiplexer 100 is located in the studio of each broadcast station, andthe transmitter 200 is located in a site placed at a predetermineddistance from the studio. The transmitter 200 may be located in aplurality of different locations. Also, for example, the plurality oftransmitters may share the same frequency. And, in this case, theplurality of transmitters receives the same signal. Accordingly, in thereceiving system, a channel equalizer may compensate signal distortion,which is caused by a reflected wave, so as to recover the originalsignal. In another example, the plurality of transmitters may havedifferent frequencies with respect to the same channel.

A variety of methods may be used for data communication each of thetransmitters, which are located in remote positions, and the servicemultiplexer. For example, an interface standard such as a synchronousserial interface for transport of MPEG-2 data (SMPTE-310M). In theSMPTE-310M interface standard, a constant data rate is decided as anoutput data rate of the service multiplexer. For example, in case of the8VSB mode, the output data rate is 19.39 Mbps, and, in case of the 16VSBmode, the output data rate is 38.78 Mbps. Furthermore, in theconventional 8VSB mode transmitting system, a transport stream (TS)packet having a data rate of approximately 19.39 Mbps may be transmittedthrough a single physical channel. Also, in the transmitting systemaccording to the present invention provided with backward compatibilitywith the conventional transmitting system, additional encoding isperformed on the mobile service data. Thereafter, the additionallyencoded mobile service data are multiplexed with the main service datato a TS packet form, which is then transmitted. At this point, the datarate of the multiplexed TS packet is approximately 19.39 Mbps.

At this point, the service multiplexer 100 receives at least one type ofmobile service data and program specific information (PSI)/program andsystem information protocol (PSIP) table data for each mobile serviceand encapsulates the received data to each transport stream (TS) packet.Also, the service multiplexer 100 receives at least one type of mainservice data and PSI/PSIP table data for each main service so as toencapsulate the received data to a TS packet. Subsequently, the TSpackets are multiplexed according to a predetermined multiplexing ruleand outputs the multiplexed packets to the transmitter 200.

Service Multiplexer

FIG. 2 illustrates a block diagram showing an example of the servicemultiplexer. The service multiplexer includes a controller 110 forcontrolling the overall operations of the service multiplexer, aPSI/PSIP generator 120 for the main service, a PSI/PSIP generator 130for the mobile service, a null packet generator 140, a mobile servicemultiplexer 150, and a transport multiplexer 160. The transportmultiplexer 160 may include a main service multiplexer 161 and atransport stream (TS) packet multiplexer 162. Referring to FIG. 2, atleast one type of compression encoded main service data and the PSI/PSIPtable data generated from the PSI/PSIP generator 120 for the mainservice are inputted to the main service multiplexer 161 of thetransport multiplexer 160. The main service multiplexer 161 encapsulateseach of the inputted main service data and PSI/PSIP table data to MPEG-2TS packet forms. Then, the MPEG-2 TS packets are multiplexed andoutputted to the TS packet multiplexer 162. Herein, the data packetbeing outputted from the main service multiplexer 161 will be referredto as a main service data packet for simplicity.

Thereafter, at least one type of the compression encoded mobile servicedata and the PSI/PSIP table data generated from the PSI/PSIP generator130 for the mobile service are inputted to the mobile servicemultiplexer 150. The mobile service multiplexer 150 encapsulates each ofthe inputted mobile service data and PSI/PSIP table data to MPEG-2 TSpacket forms. Then, the MPEG-2 TS packets are multiplexed and outputtedto the TS packet multiplexer 162. Herein, the data packet beingoutputted from the mobile service multiplexer 150 will be referred to asa mobile service data packet for simplicity. At this point, thetransmitter 200 requires identification information in order to identifyand process the main service data packet and the mobile service datapacket. Herein, the identification information may use valuespre-decided in accordance with an agreement between the transmittingsystem and the receiving system, or may be configured of a separate setof data, or may modify predetermined location value with in thecorresponding data packet. As an example of the present invention, adifferent packet identifier (PID) may be assigned to identify each ofthe main service data packet and the mobile service data packet.

In another example, by modifying a synchronization data byte within aheader of the mobile service data, the service data packet may beidentified by using the synchronization data byte value of thecorresponding service data packet. For example, the synchronization byteof the main service data packet directly outputs the value decided bythe ISO/IEC13818-1 standard (i.e., 0x47) without any modification. Thesynchronization byte of the mobile service data packet modifies andoutputs the value, thereby identifying the main service data packet andthe mobile service data packet. Conversely, the synchronization byte ofthe main service data packet is modified and outputted, whereas thesynchronization byte of the mobile service data packet is directlyoutputted without being modified, thereby enabling the main service datapacket and the mobile service data packet to be identified.

A plurality of methods may be applied in the method of modifying thesynchronization byte. For example, each bit of the synchronization bytemay be inversed, or only a portion of the synchronization byte may beinversed. As described above, any type of identification information maybe used to identify the main service data packet and the mobile servicedata packet. Therefore, the scope of the present invention is notlimited only to the example set forth in the description of the presentinvention.

Meanwhile, a transport multiplexer used in the conventional digitalbroadcasting system may be used as the transport multiplexer 160according to the present invention. More specifically, in order tomultiplex the mobile service data and the main service data and totransmit the multiplexed data, the data rate of the main service islimited to a data rate of (19.39−K) Mbps. Then, K Mbps, whichcorresponds to the remaining data rate, is assigned as the data rate ofthe mobile service. Thus, the transport multiplexer which is alreadybeing used may be used as it is without any modification. Herein, thetransport multiplexer 160 multiplexes the main service data packet beingoutputted from the main service multiplexer 161 and the mobile servicedata packet being outputted from the mobile service multiplexer 150.Thereafter, the transport multiplexer 160 transmits the multiplexed datapackets to the transmitter 200.

However, in some cases, the output data rate of the mobile servicemultiplexer 150 may not be equal to K Mbps. In this case, the mobileservice multiplexer 150 multiplexes and outputs null data packetsgenerated from the null packet generator 140 so that the output datarate can reach K Mbps. More specifically, in order to match the outputdata rate of the mobile service multiplexer 150 to a constant data rate,the null packet generator 140 generates null data packets, which arethen outputted to the mobile service multiplexer 150. For example, whenthe service multiplexer 100 assigns K Mbps of the 19.39 Mbps to themobile service data, and when the remaining (19.39-K) Mbps is,therefore, assigned to the main service data, the data rate of themobile service data that are multiplexed by the service multiplexer 100actually becomes lower than K Mbps. This is because, in case of themobile service data, the pre-processor of the transmitting systemperforms additional encoding, thereby increasing the amount of data.Eventually, the data rate of the mobile service data, which may betransmitted from the service multiplexer 100, becomes smaller than KMbps.

For example, since the pre-processor of the transmitter performs anencoding process on the mobile service data at a coding rate of at least1/2, the amount of the data outputted from the pre-processor isincreased to more than twice the amount of the data initially inputtedto the pre-processor. Therefore, the sum of the data rate of the mainservice data and the data rate of the mobile service data, both beingmultiplexed by the service multiplexer 100, becomes either equal to orsmaller than 19.39 Mbps. Therefore, in order to match the data rate ofthe data that are finally outputted from the service multiplexer 100 toa constant data rate (e.g., 19.39 Mbps), an amount of null data packetscorresponding to the amount of lacking data rate is generated from thenull packet generator 140 and outputted to the mobile servicemultiplexer 150.

Accordingly, the mobile service multiplexer 150 encapsulates each of themobile service data and the PSI/PSIP table data that are being inputtedto a MPEG-2 TS packet form. Then, the above-described TS packets aremultiplexed with the null data packets and, then, outputted to the TSpacket multiplexer 162. Thereafter, the TS packet multiplexer 162multiplexes the main service data packet being outputted from the mainservice multiplexer 161 and the mobile service data packet beingoutputted from the mobile service multiplexer 150 and transmits themultiplexed data packets to the transmitter 200 at a data rate of 19.39Mbps.

According to an embodiment of the present invention, the mobile servicemultiplexer 150 receives the null data packets. However, this is merelyexemplary and does not limit the scope of the present invention. Inother words, according to another embodiment of the present invention,the TS packet multiplexer 162 may receive the null data packets, so asto match the data rate of the finally outputted data to a constant datarate. Herein, the output path and multiplexing rule of the null datapacket is controlled by the controller 110. The controller 110 controlsthe multiplexing processed performed by the mobile service multiplexer150, the main service multiplexer 161 of the transport multiplexer 160,and the TS packet multiplexer 162, and also controls the null datapacket generation of the null packet generator 140. At this point, thetransmitter 200 discards the null data packets transmitted from theservice multiplexer 100 instead of transmitting the null data packets.

Further, in order to allow the transmitter 200 to discard the null datapackets transmitted from the service multiplexer 100 instead oftransmitting them, identification information for identifying the nulldata packet is required. Herein, the identification information may usevalues pre-decided in accordance with an agreement between thetransmitting system and the receiving system. For example, the value ofthe synchronization byte within the header of the null data packet maybe modified so as to be used as the identification information.Alternatively, a transport_error_indicator flag may also be used as theidentification information.

In the description of the present invention, an example of using thetransport_error_indicator flag as the identification information will begiven to describe an embodiment of the present invention. In this case,the transport_error_indicator flag of the null data packet is set to‘1’, and the transport_error_indicator flag of the remaining datapackets are reset to ‘0’, so as to identify the null data packet. Morespecifically, when the null packet generator 140 generates the null datapackets, if the transport_error_indicator flag from the header field ofthe null data packet is set to ‘1’ and then transmitted, the null datapacket may be identified and, therefore, be discarded. In the presentinvention, any type of identification information for identifying thenull data packets may be used. Therefore, the scope of the presentinvention is not limited only to the examples set forth in thedescription of the present invention.

According to another embodiment of the present invention, a transmissionparameter may be included in at least a portion of the null data packet,or at least one table or an operations and maintenance (OM) packet (orOMP) of the PSI/PSIP table for the mobile service. In this case, thetransmitter 200 extracts the transmission parameter and outputs theextracted transmission parameter to the corresponding block and alsotransmits the extracted parameter to the receiving system if required.More specifically, a packet referred to as an OMP is defined for thepurpose of operating and managing the transmitting system. For example,the OMP is configured in accordance with the MPEG-2 TS packet format,and the corresponding PID is given the value of 0x1FFA. The OMP isconfigured of a 4-byte header and a 184-byte payload. Herein, among the184 bytes, the first byte corresponds to an OM_type field, whichindicates the type of the OM packet.

In the present invention, the transmission parameter may be transmittedin the form of an OMP. And, in this case, among the values of thereserved fields within the OM_type field, a pre-arranged value is used,thereby indicating that the transmission parameter is being transmittedto the transmitter 200 in the form of an OMP. More specifically, thetransmitter 200 may find (or identify) the OMP by referring to the PID.Also, by parsing the OM_type field within the OMP, the transmitter 200can verify whether a transmission parameter is included after theOM_type field of the corresponding packet. The transmission parametercorresponds to supplemental data required for processing mobile servicedata from the transmitting system and the receiving system.

Herein, the transmission parameter may include data group information,region information within the data group, RS frame information, superframe information, burst information, turbo code information, and RScode information. The burst information may include burst sizeinformation, burst period information, and time information to nextburst. The burst period signifies the period at which the bursttransmitting the same mobile service is repeated. The data groupincludes a plurality of mobile service data packets, and a plurality ofsuch data groups is gathered (or grouped) to form a burst. A burstsection signifies the beginning of a current burst to the beginning of anext burst. Herein, the burst section is classified as a section thatincludes the data group (also referred to as a burst-on section), and asection that does not include the data group (also referred to as aburst-off section). A burst-on section is configured of a plurality offields, wherein one field includes one data group.

The transmission parameter may also include information on how signalsof a symbol domain are encoded in order to transmit the mobile servicedata, and multiplexing information on how the main service data and themobile service data or various types of mobile service data aremultiplexed. The information included in the transmission parameter ismerely exemplary to facilitate the understanding of the presentinvention. And, the adding and deleting of the information included inthe transmission parameter may be easily modified and changed by anyoneskilled in the art. Therefore, the present invention is not limited tothe examples proposed in the description set forth herein. Furthermore,the transmission parameters may be provided from the service multiplexer100 to the transmitter 200. Alternatively, the transmission parametersmay also be set up by an internal controller (not shown) within thetransmitter 200 or received from an external source.

Transmitter

FIG. 3 illustrates a block diagram showing an example of the transmitter200 according to an embodiment of the present invention. Herein, thetransmitter 200 includes a demultiplexer 210, a packet jitter mitigator220, a pre-processor 230, a packet multiplexer 240, a post-processor250, a synchronization (sync) multiplexer 260, and a transmission unit270. Herein, when a data packet is received from the service multiplexer100, the demultiplexer 210 should identify whether the received datapacket corresponds to a main service data packet, a mobile service datapacket, or a null data packet. For example, the demultiplexer 210 usesthe PID within the received data packet so as to identify the mainservice data packet and the mobile service data packet. Then, thedemultiplexer 210 uses a transport_error_indicator field to identify thenull data packet. The main service data packet identified by thedemultiplexer 210 is outputted to the packet jitter mitigator 220, themobile service data packet is outputted to the pre-processor 230, andthe null data packet is discarded. If a transmission parameter isincluded in the null data packet, then the transmission parameter isfirst extracted and outputted to the corresponding block. Thereafter,the null data packet is discarded.

The pre-processor 230 performs an additional encoding process of themobile service data included in the service data packet, which isdemultiplexed and outputted from the demultiplexer 210. Thepre-processor 230 also performs a process of configuring a data group sothat the data group may be positioned at a specific place in accordancewith the purpose of the data, which are to be transmitted on atransmission frame. This is to enable the mobile service data to respondswiftly and strongly against noise and channel changes. Thepre-processor 230 may also refer to the transmission parameter whenperforming the additional encoding process. Also, the pre-processor 230groups a plurality of mobile service data packets to configure a datagroup. Thereafter, known data, mobile service data, RS parity data, andMPEG header are allocated to pre-determined areas within the data group.

Pre-Processor within Transmitter

FIG. 4 illustrates a block diagram showing an example of thepre-processor 230 according to the present invention. The pre-processor230 includes a data randomizer 301, a RS frame encoder 302, a blockprocessor 303, a group formatter 304, a data deinterleaver 305, a packetformatter 306. The data randomizer 301 within the above-describedpre-processor 230 randomizes the mobile service data packet includingthe mobile service data that is inputted through the demultiplexer 210.Then, the data randomizer 301 outputs the randomized mobile service datapacket to the RS frame encoder 302. At this point, since the datarandomizer 301 performs the randomizing process on the mobile servicedata, the randomizing process that is to be performed by the datarandomizer 251 of the post-processor 250 on the mobile service data maybe omitted. The data randomizer 301 may also discard the synchronizationbyte within the mobile service data packet and perform the randomizingprocess. This is an option that may be chosen by the system designer. Inthe example given in the present invention, the randomizing process isperformed without discarding the synchronization byte within the mobileservice data packet.

The RS frame encoder 302 groups a plurality of mobile thesynchronization byte within the mobile service data packets that israndomized and inputted, so as to create a RS frame. Then, the RS frameencoder 302 performs at least one of an error correction encodingprocess and an error detection encoding process in RS frame units.Accordingly, robustness may be provided to the mobile service data,thereby scattering group error that may occur during changes in afrequency environment, thereby enabling the mobile service data torespond to the frequency environment, which is extremely vulnerable andliable to frequent changes. Also, the RS frame encoder 302 groups aplurality of RS frame so as to create a super frame, thereby performinga row permutation process in super frame units. The row permutationprocess may also be referred to as a row interleaving process.Hereinafter, the process will be referred to as row permutation forsimplicity.

More specifically, when the RS frame encoder 302 performs the process ofpermuting each row of the super frame in accordance with apre-determined rule, the position of the rows within the super framebefore and after the row permutation process is changed. If the rowpermutation process is performed by super frame units, and even thoughthe section having a plurality of errors occurring therein becomes verylong, and even though the number of errors included in the RS frame,which is to be decoded, exceeds the extent of being able to becorrected, the errors become dispersed within the entire super frame.Thus, the decoding ability is even more enhanced as compared to a singleRS frame.

At this point, as an example of the present invention, RS-encoding isapplied for the error correction encoding process, and a cyclicredundancy check (CRC) encoding is applied for the error detectionprocess. When performing the RS-encoding, parity data that are used forthe error correction are generated. And, when performing the CRCencoding, CRC data that are used for the error detection are generated.The RS encoding is one of forward error correction (FEC) methods. TheFEC corresponds to a technique for compensating errors that occur duringthe transmission process. The CRC data generated by CRC encoding may beused for indicating whether or not the mobile service data have beendamaged by the errors while being transmitted through the channel. Inthe present invention, a variety of error detection coding methods otherthan the CRC encoding method may be used, or the error correction codingmethod may be used to enhance the overall error correction ability ofthe receiving system. Herein, the RS frame encoder 302 refers to apre-determined transmission parameter and/or the transmission parameterprovided from the service multiplexer 100 so as to perform operationsincluding RS frame configuration, RS encoding, CRC encoding, super frameconfiguration, and row permutation in super frame units.

Pre-Processor within RS Frame Encoder

FIG. 5( a) to FIG. 5( e) illustrate error correction encoding and errordetection encoding processed according to an embodiment of the presentinvention. More specifically, the RS frame encoder 302 first divides theinputted mobile service data bytes to units of a predetermined length.The predetermined length is decided by the system designer. And, in theexample of the present invention, the predetermined length is equal to187 bytes, and, therefore, the 187-byte unit will be referred to as apacket for simplicity. For example, when the mobile service data thatare being inputted, as shown in FIG. 5( a), correspond to a MPEGtransport packet stream configured of 188-byte units, the firstsynchronization byte is removed, as shown in FIG. 5( b), so as toconfigure a 187-byte unit. Herein, the synchronization byte is removedbecause each mobile service data packet has the same value.

Herein, the process of removing the synchronization byte may beperformed during a randomizing process of the data randomizer 301 in anearlier process. In this case, the process of the removing thesynchronization byte by the RS frame encoder 302 may be omitted.Moreover, when adding synchronization bytes from the receiving system,the process may be performed by the data derandomizer instead of the RSframe decoder. Therefore, if a removable fixed byte (e.g.,synchronization byte) does not exist within the mobile service datapacket that is being inputted to the RS frame encoder 302, or if themobile service data that are being inputted are not configured in apacket format, the mobile service data that are being inputted aredivided into 187-byte units, thereby configuring a packet for each187-byte unit.

Subsequently, as shown in FIG. 5( c), N number of packets configured of187 bytes is grouped to configure a RS frame. At this point, the RSframe is configured as a RS frame having the size of N(row)*187(column)bytes, in which 187-byte packets are sequentially inputted in a rowdirection. In order to simplify the description of the presentinvention, the RS frame configured as described above will also bereferred to as a first RS frame. More specifically, only pure mobileservice data are included in the first RS frame, which is the same asthe structure configured of 187 N-byte rows. Thereafter, the mobileservice data within the RS frame are divided into an equal size. Then,when the divided mobile service data are transmitted in the same orderas the input order for configuring the RS frame, and when one or moreerrors have occurred at a particular point during thetransmitting/receiving process, the errors are clustered (or gathered)within the RS frame as well. In this case, the receiving system uses aRS erasure decoding method when performing error correction decoding,thereby enhancing the error correction ability. At this point, the Nnumber of columns within the N number of RS frame includes 187 bytes, asshown in FIG. 5( c).

In this case, a (Nc,Kc)−RS encoding process is performed on each column,so as to generate Nc−Kc(=P) number of parity bytes. Then, the newlygenerated P number of parity bytes is added after the very last byte ofthe corresponding column, thereby creating a column of (187+P) bytes.Herein, as shown in FIG. 5( c), Kc is equal to 187 (i.e., Kc=187), andNc is equal to 187+P (i.e., Nc=187+P). For example, when P is equal to48, (235,187)−RS encoding process is performed so as to create a columnof 235 bytes. When such RS encoding process is performed on all N numberof columns, as shown in FIG. 5( c), a RS frame having the size ofN(row)*(187+P)(column) bytes may be created, as shown in FIG. 5( d). Inorder to simplify the description of the present invention, the RS framehaving the RS parity inserted therein will be referred to as s second RSframe. More specifically, the second RS frame having the structure of(187+P) rows configured of N bytes may be configured.

As shown in FIG. 5( c) or FIG. 5( d), each row of the RS frame isconfigured of N bytes. However, depending upon channel conditionsbetween the transmitting system and the receiving system, error may beincluded in the RS frame. When errors occur as described above, CRC data(or CRC code or CRC checksum) may be used on each row unit in order toverify whether error exists in each row unit. The RS frame encoder 302may perform CRC encoding on the mobile service data being RS encoded soas to create (or generate) the CRC data. The CRC data being generated byCRC encoding may be used to indicate whether the mobile service datahave been damaged while being transmitted through the channel.

The present invention may also use different error detection encodingmethods other than the CRC encoding method. Alternatively, the presentinvention may use the error correction encoding method to enhance theoverall error correction ability of the receiving system. FIG. 5( e)illustrates an example of using a 2-byte (i.e., 16-bit) CRC checksum asthe CRC data. Herein, a 2-byte CRC checksum is generated for N number ofbytes of each row, thereby adding the 2-byte CRC checksum at the end ofthe N number of bytes. Thus, each row is expanded to (N+2) number ofbytes. Equation 1 below corresponds to an exemplary equation forgenerating a 2-byte CRC checksum for each row being configured of Nnumber of bytes.

g(x)=x ¹⁶ +x ¹² +x ⁵+1  Equation 1

The process of adding a 2-byte checksum in each row is only exemplary.Therefore, the present invention is not limited only to the exampleproposed in the description set forth herein. In order to simplify theunderstanding of the present invention, the RS frame having the RSparity and CRC checksum added therein will hereinafter be referred to asa third RS frame. More specifically, the third RS frame corresponds to(187+P) number of rows each configured of (N+2) number of bytes. Asdescribed above, when the process of RS encoding and CRC encoding arecompleted, the (N*187)-byte RS frame is expanded to a (N+2)(187+P)-byteRS frame. Furthermore, the RS frame that is expanded, as shown in FIG.5( e), is inputted to the block processor 303.

As described above, the mobile service data encoded by the RS frameencoder 302 are inputted to the block processor 303. The block processor303 then encodes the inputted mobile service data at a coding rate ofG/H (wherein, G is smaller than H (i.e., G<H)) and then outputted to thegroup formatter 304. More specifically, the block processor 303 dividesthe mobile service data being inputted in byte units into bit units.Then, the G number of bits is encoded to H number of bits. Thereafter,the encoded bits are converted back to byte units and then outputted.For example, if 1 bit of the input data is coded to 2 bits andoutputted, then G is equal to 1 and H is equal to 2 (i.e., G=1 and H=2).Alternatively, if 1 bit of the input data is coded to 4 bits andoutputted, then G is equal to 1 and H is equal to 4 (i.e., G=1 and H=4).Hereinafter, the former coding rate will be referred to as a coding rateof 1/2 (1/2-rate coding), and the latter coding rate will be referred toas a coding rate of 1/4 (1/4-rate coding), for simplicity.

Herein, when using the 1/4 coding rate, the coding efficiency is greaterthan when using the 1/2 coding rate, and may, therefore, provide greaterand enhanced error correction ability. For such reason, when it isassumed that the data encoded at a 1/4 coding rate in the groupformatter 304, which is located near the end portion of the system, areallocated to an area in which the receiving performance may bedeteriorated, and that the data encoded at a 1/2 coding rate areallocated to an area having excellent receiving performance, thedifference in performance may be reduced. At this point, the blockprocessor 303 may also receive signaling information includingtransmission parameters. Herein, the signaling information may also beprocessed with either 1/2-rate coding or 1/4-rate coding as in the stepof processing mobile service data. Thereafter, the signaling informationis also considered the same as the mobile service data and processedaccordingly.

Meanwhile, the group formatter inserts mobile service data that areoutputted from the block processor 303 in corresponding areas within adata group, which is configured in accordance with a pre-defined rule.Also, with respect to the data deinterleaving process, each place holderor known data (or known data place holders) are also inserted incorresponding areas within the data group. At this point, the data groupmay be divided into at least one hierarchical area. Herein, the type ofmobile service data being inserted in each area may vary depending uponthe characteristics of each hierarchical area. Additionally, each areamay, for example, be divided based upon the receiving performance withinthe data group. Furthermore, one data group may be configured to includea set of field synchronization data.

In an example given in the present invention, a data group is dividedinto A, B, and C regions in a data configuration prior to datadeinterleaving. At this point, the group formatter 304 allocates themobile service data, which are inputted after being RS encoded and blockencoded, to each of the corresponding regions by referring to thetransmission parameter. FIG. 6A illustrates an alignment of data afterbeing data interleaved and identified, and FIG. 6B illustrates analignment of data before being data interleaved and identified. Morespecifically, a data structure identical to that shown in FIG. 6A istransmitted to a receiving system. Also, the data group configured tohave the same structure as the data structure shown in FIG. 6A isinputted to the data deinterleaver 305.

As described above, FIG. 6A illustrates a data structure prior to datadeinterleaving that is divided into 3 regions, such as region A, regionB, and region C. Also, in the present invention, each of the regions Ato C is further divided into a plurality of regions. Referring to FIG.6A, region A is divided into 5 regions (A1 to A5), region B is dividedinto 2 regions (B1 and B2), and region C is divided into 3 regions (C1to C3). Herein, regions A to C are identified as regions having similarreceiving performances within the data group. Herein, the type of mobileservice data, which are inputted, may also vary depending upon thecharacteristic of each region.

In the example of the present invention, the data structure is dividedinto regions A to C based upon the level of interference of the mainservice data. Herein, the data group is divided into a plurality ofregions to be used for different purposes. More specifically, a regionof the main service data having no interference or a very lowinterference level may be considered to have a more resistant (orstronger) receiving performance as compared to regions having higherinterference levels. Additionally, when using a system inserting andtransmitting known data in the data group, and when consecutively longknown data are to be periodically inserted in the mobile service data,the known data having a predetermined length may be periodicallyinserted in the region having no interference from the main service data(e.g., region A). However, due to interference from the main servicedata, it is difficult to periodically insert known data and also toinsert consecutively long known data to a region having interferencefrom the main service data (e.g., region B and region C).

Hereinafter, examples of allocating data to region A (A1 to A5), regionB (B1 and B2), and region C (C1 to C3) will now be described in detailwith reference to FIG. 6A. The data group size, the number ofhierarchically divided regions within the data group and the size ofeach region, and the number of mobile service data bytes that can beinserted in each hierarchically divided region of FIG. 6A are merelyexamples given to facilitate the understanding of the present invention.Herein, the group formatter 304 creates a data group including places inwhich field synchronization data bytes are to be inserted, so as tocreate the data group that will hereinafter be described in detail.

More specifically, region A is a region within the data group in which along known data sequence may be periodically inserted, and in whichincludes regions wherein the main service data are not mixed (e.g., A1to A5). Also, region A includes a region (e.g., A1) located between afield synchronization region and the region in which the first knowndata sequence is to be inserted. The field synchronization region hasthe length of one segment (i.e., 832 symbols) existing in an ATSCsystem.

For example, referring to FIG. 6A, 2428 bytes of the mobile service datamay be inserted in region A1, 2580 bytes may be inserted in region A2,2772 bytes may be inserted in region A3, 2472 bytes may be inserted inregion A4, and 2772 bytes may be inserted in region A5. Herein, trellisinitialization data or known data, MPEG header, and RS parity are notincluded in the mobile service data. As described above, when region Aincludes a known data sequence at both ends, the receiving system useschannel information that can obtain known data or field synchronizationdata, so as to perform equalization, thereby providing enforcedequalization performance.

Also, region B includes a region located within 8 segments at thebeginning of a field synchronization region within the data group(chronologically placed before region A1) (e.g., region B1), and aregion located within 8 segments behind the very last known datasequence which is inserted in the data group (e.g., region B2). Forexample, 930 bytes of the mobile service data may be inserted in theregion B1, and 1350 bytes may be inserted in region B2. Similarly,trellis initialization data or known data, MPEG header, and RS parityare not included in the mobile service data. In case of region B, thereceiving system may perform equalization by using channel informationobtained from the field synchronization region. Alternatively, thereceiving system may also perform equalization by using channelinformation that may be obtained from the last known data sequence,thereby enabling the system to respond to the channel changes.

Region C includes a region located within 30 segments including andpreceding the 9^(th) segment of the field synchronization region(chronologically located before region A) (e.g., region C1), a regionlocated within 12 segments including and following the 9^(th) segment ofthe very last known data sequence within the data group (chronologicallylocated after region A) (e.g., region C2), and a region located in 32segments after the region C2 (e.g., region C3). For example, 1272 bytesof the mobile service data may be inserted in the region C1, 1560 bytesmay be inserted in region C2, and 1312 bytes may be inserted in regionC3. Similarly, trellis initialization data or known data, MPEG header,and RS parity are not included in the mobile service data. Herein,region C (e.g., region C1) is located chronologically earlier than (orbefore) region A.

Since region C (e.g., region C1) is located further apart from the fieldsynchronization region which corresponds to the closest known dataregion, the receiving system may use the channel information obtainedfrom the field synchronization data when performing channelequalization. Alternatively, the receiving system may also use the mostrecent channel information of a previous data group. Furthermore, inregion C (e.g., region C2 and region C3) located before region A, thereceiving system may use the channel information obtained from the lastknown data sequence to perform equalization. However, when the channelsare subject to fast and frequent changes, the equalization may not beperformed perfectly. Therefore, the equalization performance of region Cmay be deteriorated as compared to that of region B.

When it is assumed that the data group is allocated with a plurality ofhierarchically divided regions, as described above, the block processor303 may encode the mobile service data, which are to be inserted to eachregion based upon the characteristic of each hierarchical region, at adifferent coding rate. For example, the block processor 303 may encodethe mobile service data, which are to be inserted in regions A1 to A5 ofregion A, at a coding rate of 1/2. Then, the group formatter 304 mayinsert the 1/2-rate encoded mobile service data to regions A1 to A5.

The block processor 303 may encode the mobile service data, which are tobe inserted in regions B1 and B2 of region B, at a coding rate of 1/4having higher error correction ability as compared to the 1/2-codingrate. Then, the group formatter 304 inserts the 1/4-rate coded mobileservice data in region B1 and region B2. Furthermore, the blockprocessor 303 may encode the mobile service data, which are to beinserted in regions C1 to C3 of region C, at a coding rate of 1/4 or acoding rate having higher error correction ability than the 1/4-codingrate. Then, the group formatter 304 may either insert the encoded mobileservice data to regions C1 to C3, as described above, or leave the datain a reserved region for future usage.

In addition, the group formatter 304 also inserts supplemental data,such as signaling information that notifies the overall transmissioninformation, other than the mobile service data in the data group. Also,apart from the encoded mobile service data outputted from the blockprocessor 303, the group formatter 304 also inserts MPEG header placeholders, non-systematic RS parity place holders, main service data placeholders, which are related to data deinterleaving in a later process, asshown in FIG. 6A. Herein, the main service data place holders areinserted because the mobile service data bytes and the main service databytes are alternately mixed with one another in regions B and C basedupon the input of the data deinterleaver, as shown in FIG. 6A. Forexample, based upon the data outputted after data deinterleaving, theplace holder for the MPEG header may be allocated at the very beginningof each packet.

Furthermore, the group formatter 304 either inserts known data generatedin accordance with a pre-determined method or inserts known data placeholders for inserting the known data in a later process. Additionally,place holders for initializing the trellis encoding module 256 are alsoinserted in the corresponding regions. For example, the initializationdata place holders may be inserted in the beginning of the known datasequence. Herein, the size of the mobile service data that can beinserted in a data group may vary in accordance with the sizes of thetrellis initialization place holders or known data (or known data placeholders), MPEG header place holders, and RS parity place holders.

The output of the group formatter 304 is inputted to the datadeinterleaver 305. And, the data deinterleaver 305 deinterleaves data byperforming an inverse process of the data interleaver on the data andplace holders within the data group, which are then outputted to thepacket formatter 306. More specifically, when the data and place holderswithin the data group configured, as shown in FIG. 6A, are deinterleavedby the data deinterleaver 305, the data group being outputted to thepacket formatter 306 is configured to have the structure shown in FIG.6B.

The packet formatter 306 removes the main service data place holders andthe RS parity place holders that were allocated for the deinterleavingprocess from the deinterleaved data being inputted. Then, the packetformatter 306 groups the remaining portion and replaces the 4-byte MPEGheader place holder with an MPEG header having a null packet PID (or anunused PID from the main service data packet). Also, when the groupformatter 304 inserts known data place holders, the packet formatter 306may insert actual known data in the known data place holders, or maydirectly output the known data place holders without any modification inorder to make replacement insertion in a later process. Thereafter, thepacket formatter 306 identifies the data within the packet-formatteddata group, as described above, as a 188-byte unit mobile service datapacket (i.e., MPEG TS packet), which is then provided to the packetmultiplexer 240.

The packet multiplexer 240 multiplexes the mobile service data packetoutputted from the pre-processor 230 and the main service data packetoutputted from the packet jitter mitigator 220 in accordance with apre-defined multiplexing method. Then, the packet multiplexer 240outputs the multiplexed data packets to the data randomizer 251 of thepost-processor 250. Herein, the multiplexing method may vary inaccordance with various variables of the system design. One of themultiplexing methods of the packet formatter 240 consists of providing aburst section along a time axis, and, then, transmitting a plurality ofdata groups during a burst-on section within the burst section, andtransmitting only the main service data during the burst-off sectionwithin the burst section. Herein, the burst section indicates thesection starting from the beginning of the current burst until thebeginning of the next burst.

At this point, the main service data may be transmitted during theburst-on section. The packet multiplexer 240 refers to the transmissionparameter, such as information on the burst size or the burst period, soas to be informed of the number of data groups and the period of thedata groups included in a single burst. Herein, the mobile service dataand the main service data may co-exist in the burst-on section, and onlythe main service data may exist in the burst-off section. Therefore, amain data service section transmitting the main service data may existin both burst-on and burst-off sections. At this point, the main dataservice section within the burst-on section and the number of main dataservice packets included in the burst-off section may either bedifferent from one another or be the same.

When the mobile service data are transmitted in a burst structure, inthe receiving system receiving only the mobile service data turns thepower on only during the burst section, thereby receiving thecorresponding data. Alternatively, in the section transmitting only themain service data, the power is turned off so that the main service dataare not received in this section. Thus, the power consumption of thereceiving system may be reduced.

Detailed Embodiments of the RS Frame Structure and Packet Multiplexing

Hereinafter, detailed embodiments of the pre-processor 230 and thepacket multiplexer 240 will now be described. According to an embodimentof the present invention, the N value corresponding to the length of arow, which is included in the RS frame that is configured by the RSframe encoder 302, is set to 538. Accordingly, the RS frame encoder 302receives 538 transport stream (TS) packets so as to configure a first RSframe having the size of 538*187 bytes. Thereafter, as described above,the first RS frame is processed with a (235,187)−RS encoding process soas to configure a second RS frame having the size of 538*235 bytes.Finally, the second RS frame is processed with generating a 16-bitchecksum so as to configure a third RS frame having the sizes of540*235.

Meanwhile, as shown in FIG. 6A, the sum of the number of bytes ofregions A1 to A5 of region A, in which 1/2-rate encoded mobile servicedata are to be inserted, among the plurality of regions within the datagroup is equal to 13024 bytes (=2428+2580+2772+2472+2772 bytes). Herein,the number of byte prior to performing the 1/2-rate encoding process isequal to 6512 (=13024/2). On the other hand, the sum of the number ofbytes of regions B1 and B2 of region B, in which 1/4-rate encoded mobileservice data are to be inserted, among the plurality of regions withinthe data group is equal to 2280 bytes (=930+1350 bytes). Herein, thenumber of byte prior to performing the 1/4-rate encoding process isequal to 570 (=2280/4).

In other words, when 7082 bytes of mobile service data are inputted tothe block processor 303, 6512 byte are expanded to 13024 bytes by being1/2-rate encoded, and 570 bytes are expanded to 2280 bytes by being1/4-rate encoded. Thereafter, the block processor 303 inserts the mobileservice data expanded to 13024 bytes in regions A1 to A5 of region Aand, also, inserts the mobile service data expanded to 2280 bytes inregions B1 and B2 of region B. Herein, the 7082 bytes of mobile servicedata being inputted to the block processor 303 may be divided into anoutput of the RS frame encoder 302 and signaling information. In thepresent invention, among the 7082 bytes of mobile service data, 7050bytes correspond to the output of the RS frame encoder 302, and theremaining 32 bytes correspond to the signaling information data. Then,1/2-rate encoding or 1/4-rate encoding is performed on the correspondingdata bytes.

Meanwhile, a RS frame being processed with RS encoding and CRC encodingfrom the RS frame encoder 302 is configured of 540*235 bytes, in otherwords, 126900 bytes. The 126900 bytes are divided by 7050-byte unitsalong the time axis, so as to produce 18 7050-byte units. Thereafter, a32-byte unit of signaling information data is added to the 7050-byteunit mobile service data being outputted from the RS frame encoder 302.Subsequently, the RS frame encoder 302 performs 1/2-rate encoding or1/4-rate encoding on the corresponding data bytes, which are thenoutputted to the group formatter 304. Accordingly, the group formatter304 inserts the 1/2-rate encoded data in region A and the 1/4-rateencoded data in region B.

The process of deciding an N value that is required for configuring theRS frame from the RS frame encoder 302 will now be described in detail.More specifically, the size of the final RS frame (i.e., the third RSframe), which is RS encoded and CRC encoded from the RS frame encoder302, which corresponds to (N+2)235 bytes should be allocated to X numberof groups, wherein X is an integer. Herein, in a single data group, 7050data bytes prior to being encoded are allocated. Therefore, if the(N+2)*235 bytes are set to be the exact multiple of 7050(=30*235), theoutput data of the RS frame encoder 302 may be efficiently allocated tothe data group. According to an embodiment of the present invention, thevalue of N is decided so that (N+2) becomes a multiple of 30. Forexample, in the present invention, N is equal to 538, and (N+2)(=540)divided by 30 is equal to 18. This indicates that the mobile servicedata within one RS frame are processed with either 1/2-rate encoding or1/4-rate encoding. The encoded mobile service data are then allocated to18 data groups.

FIG. 7 illustrates a process of dividing the RS frame according to thepresent invention. More specifically, the RS frame having the size of(N+2)*235 is divided into 30*235 byte blocks. Then, the divided blocksare mapped to a single group. In other words, the data of a block havingthe size of 30*235 bytes are processed with one of a 1/2-rate encodingprocess and a 1/4-rate encoding process and are, then, inserted in adata group. Thereafter, the data group having corresponding data andplace holders inserted in each hierarchical region divided by the groupformatter 304 passes through the data deinterleaver 305 and the packetformatter 306 so as to be inputted to the packet multiplexer 240.

FIG. 8 illustrates exemplary operations of a packet multiplexer fortransmitting the data group according to the present invention. Morespecifically, the packet multiplexer 240 multiplexes a field including adata group, in which the mobile service data and main service data aremixed with one another, and a field including only the main servicedata. Thereafter, the packet multiplexer 240 outputs the multiplexedfields to the data randomizer 251. At this point, in order to transmitthe RS frame having the size of 540*235 bytes, 18 data groups should betransmitted. Herein, each data group includes field synchronizationdata, as shown in FIG. 6A. Therefore, the 18 data groups are transmittedduring 18 field sections, and the section during which the 18 datagroups are being transmitted corresponds to the burst-on section.

In each field within the burst-on section, a data group including fieldsynchronization data is multiplexed with main service data, which arethen outputted. For example, in the embodiment of the present invention,in each field within the burst-on section, a data group having the sizeof 118 segments is multiplexed with a set of main service data havingthe size of 194 segments. Referring to FIG. 8, during the burst-onsection (i.e., during the 18 field sections), a field including 18 datagroups is transmitted. Then, during the burst-off section that follows(i.e., during the 12 field sections), a field consisting only of themain service data is transmitted. Subsequently, during a subsequentburst-on section, 18 fields including 18 data groups are transmitted.And, during the following burst-off section, 12 fields consisting onlyof the main service data are transmitted.

Furthermore, in the present invention, the same type of data service maybe provided in the first burst-on section including the first 18 datagroups and in the second burst-on section including the next 18 datagroups. Alternatively, different types of data service may be providedin each burst-on section. For example, when it is assumed that differentdata service types are provided to each of the first burst-on sectionand the second burst-on section, and that the receiving system wishes toreceive only one type of data service, the receiving system turns thepower on only during the corresponding burst-on section including thedesired data service type so as to receive the corresponding 18 datafields. Then, the receiving system turns the power off during theremaining 42 field sections so as to prevent other data service typesfrom being received. Thus, the amount of power consumption of thereceiving system may be reduced. In addition, the receiving systemaccording to the present invention is advantageous in that one RS framemay be configured from the 18 data groups that are received during asingle burst-on section.

According to the present invention, the number of data groups includedin a burst-on section may vary based upon the size of the RS frame, andthe size of the RS frame varies in accordance with the value N. Morespecifically, by adjusting the value N, the number of data groups withinthe burst section may be adjusted. Herein, in an example of the presentinvention, the (235,187)−RS encoding process adjusts the value N duringa fixed state. Furthermore, the size of the mobile service data that canbe inserted in the data group may vary based upon the sizes of thetrellis initialization data or known data, the MPEG header, and the RSparity, which are inserted in the corresponding data group.

Meanwhile, since a data group including mobile service data in-betweenthe data bytes of the main service data during the packet multiplexingprocess, the shifting of the chronological position (or place) of themain service data packet becomes relative. Also, a system object decoder(i.e., MPEG decoder) for processing the main service data of thereceiving system, receives and decodes only the main service data andrecognizes the mobile service data packet as a null data packet.Therefore, when the system object decoder of the receiving systemreceives a main service data packet that is multiplexed with the datagroup, a packet jitter occurs.

At this point, since a multiple-level buffer for the video data existsin the system object decoder and the size of the buffer is relativelylarge, the packet jitter generated from the packet multiplexer 240 doesnot cause any serious problem in case of the video data. However, sincethe size of the buffer for the audio data is relatively small, thepacket jitter may cause considerable problem. More specifically, due tothe packet jitter, an overflow or underflow may occur in the buffer forthe main service data of the receiving system (e.g., the buffer for theaudio data). Therefore, the packet jitter mitigator 220 re-adjusts therelative position of the main service data packet so that the overflowor underflow does not occur in the system object decoder.

In the present invention, examples of repositioning places for the audiodata packets within the main service data in order to minimize theinfluence on the operations of the audio buffer will be described indetail. The packet jitter mitigator 220 repositions the audio datapackets in the main service data section so that the audio data packetsof the main service data can be as equally and uniformly aligned andpositioned as possible. The standard for repositioning the audio datapackets in the main service data performed by the packet jittermitigator 220 will now be described. Herein, it is assumed that thepacket jitter mitigator 220 knows the same multiplexing information asthat of the packet multiplexer 240, which is placed further behind thepacket jitter mitigator 220.

Firstly, if one audio data packet exists in the main service datasection (e.g., the main service data section positioned between two datagroups) within the burst-on section, the audio data packet is positionedat the very beginning of the main service data section. Alternatively,if two audio data packets exist in the corresponding data section, oneaudio data packet is positioned at the very beginning and the otheraudio data packet is positioned at the very end of the main service datasection. Further, if more than three audio data packets exist, one audiodata packet is positioned at the very beginning of the main service datasection, another is positioned at the very end of the main service datasection, and the remaining audio data packets are equally positionedbetween the first and last audio data packets. Secondly, during the mainservice data section placed immediately before the beginning of aburst-on section (i.e., during a burst-off section), the audio datapacket is placed at the very end of the corresponding section.

Thirdly, during a main service data section within the burst-off sectionafter the burst-on section, the audio data packet is positioned at thevery end of the main service data section. Finally, the data packetsother than audio data packets are positioned in accordance with theinputted order in vacant spaces (i.e., spaces that are not designatedfor the audio data packets). Meanwhile, when the positions of the mainservice data packets are relatively re-adjusted, associated programclock reference (PCR) values may also be modified accordingly. The PCRvalue corresponds to a time reference value for synchronizing the timeof the MPEG decoder. Herein, the PCR value is inserted in a specificregion of a TS packet and then transmitted.

In the example of the present invention, the packet jitter mitigator 220also performs the operation of modifying the PCR value. The output ofthe packet jitter mitigator 220 is inputted to the packet multiplexer240. As described above, the packet multiplexer 240 multiplexes the mainservice data packet outputted from the packet jitter mitigator 220 withthe mobile service data packet outputted from the pre-processor 230 intoa burst structure in accordance with a pre-determined multiplexing rule.Then, the packet multiplexer 240 outputs the multiplexed data packets tothe data randomizer 251 of the post-processor 250.

If the inputted data correspond to the main service data packet, thedata randomizer 251 performs the same randomizing process as that of theconventional randomizer. More specifically, the synchronization bytewithin the main service data packet is deleted. Then, the remaining 187data bytes are randomized by using a pseudo random byte generated fromthe data randomizer 251. Thereafter, the randomized data are outputtedto the RS encoder/non-systematic RS encoder 252.

On the other hand, if the inputted data correspond to the mobile servicedata packet, the data randomizer 251 may randomize only a portion of thedata packet. For example, if it is assumed that a randomizing processhas already been performed in advance on the mobile service data packetby the pre-processor 230, the data randomizer 251 deletes thesynchronization byte from the 4-byte MPEG header included in the mobileservice data packet and, then, performs the randomizing process only onthe remaining 3 data bytes of the MPEG header. Thereafter, therandomized data bytes are outputted to the RS encoder/non-systematic RSencoder 252. More specifically, the randomizing process is not performedon the remaining portion of the mobile service data excluding the MPEGheader. In other words, the remaining portion of the mobile service datapacket is directly outputted to the RS encoder/non-systematic RS encoder252 without being randomized. Also, the data randomizer 251 may or maynot perform a randomizing process on the known data (or known data placeholders) and the initialization data place holders included in themobile service data packet.

The RS encoder/non-systematic RS encoder 252 performs an RS encodingprocess on the data being randomized by the data randomizer 251 or onthe data bypassing the data randomizer 251, so as to add 20 bytes of RSparity data. Thereafter, the processed data are outputted to the datainterleaver 253. Herein, if the inputted data correspond to the mainservice data packet, the RS encoder/non-systematic RS encoder 252performs the same systematic RS encoding process as that of theconventional broadcasting system, thereby adding the 20-byte RS paritydata at the end of the 187-byte data. Alternatively, if the inputteddata correspond to the mobile service data packet, the RSencoder/non-systematic RS encoder 252 performs a non-systematic RSencoding process. At this point, the 20-byte RS parity data obtainedfrom the non-systematic RS encoding process are inserted in apre-decided parity byte place within the mobile service data packet.

The data interleaver 253 corresponds to a byte unit convolutionalinterleaver. The output of the data interleaver 253 is inputted to theparity replacer 254 and to the non-systematic RS encoder 255. Meanwhile,a process of initializing a memory within the trellis encoding module256 is primarily required in order to decide the output data of thetrellis encoding module 256, which is located after the parity replacer254, as the known data pre-defined according to an agreement between thereceiving system and the transmitting system. More specifically, thememory of the trellis encoding module 256 should first be initializedbefore the received known data sequence is trellis-encoded. At thispoint, the beginning portion of the known data sequence that is receivedcorresponds to the initialization data place holder and not to theactual known data. Herein, the initialization data place holder has beenincluded in the data by the group formatter within the pre-processor 230in an earlier process. Therefore, the process of generatinginitialization data and replacing the initialization data place holderof the corresponding memory with the generated initialization data arerequired to be performed immediately before the inputted known datasequence is trellis-encoded.

Additionally, a value of the trellis memory initialization data isdecided and generated based upon a memory status of the trellis encodingmodule 256. Further, due to the newly replaced initialization data, aprocess of newly calculating the RS parity and replacing the RS parity,which is outputted from the data interleaver 253, with the newlycalculated RS parity is required. Therefore, the non-systematic RSencoder 255 receives the mobile service data packet including theinitialization data place holders, which are to be replaced with theactual initialization data, from the data interleaver 253 and alsoreceives the initialization data from the trellis encoding module 256.

Among the inputted mobile service data packet, the initialization dataplace holders are replaced with the initialization data, and the RSparity data that are added to the mobile service data packet are removedand processed with non-systematic RS encoding. Thereafter, the new RSparity obtained by performing the non-systematic RS encoding process isoutputted to the parity replacer 255. Accordingly, the parity replacer255 selects the output of the data interleaver 253 as the data withinthe mobile service data packet, and the parity replacer 255 selects theoutput of the non-systematic RS encoder 255 as the RS parity. Theselected data are then outputted to the trellis encoding module 256.

Meanwhile, if the main service data packet is inputted or if the mobileservice data packet, which does not include any initialization dataplace holders that are to be replaced, is inputted, the parity replacer254 selects the data and RS parity that are outputted from the datainterleaver 253. Then, the parity replacer 254 directly outputs theselected data to the trellis encoding module 256 without anymodification. The trellis encoding module 256 converts the byte-unitdata to symbol units and performs a 12-way interleaving process so as totrellis-encode the received data. Thereafter, the processed data areoutputted to the synchronization multiplexer 260.

The synchronization multiplexer 260 inserts a field synchronizationsignal and a segment synchronization signal to the data outputted fromthe trellis encoding module 256 and, then, outputs the processed data tothe pilot inserter 271 of the transmission unit 270. Herein, the datahaving a pilot inserted therein by the pilot inserter 271 are modulatedby the modulator 272 in accordance with a pre-determined modulatingmethod (e.g., a VSB method). Thereafter, the modulated data aretransmitted to each receiving system though the radio frequency (RF)up-converter 273.

Block Processor

FIG. 9 illustrates a block diagram showing a structure of a blockprocessor according to the present invention. Herein, the blockprocessor includes a byte-bit converter 401, a symbol encoder 402, asymbol interleaver 403, and a symbol-byte converter 404. The byte-bitconverter 401 divides the mobile service data bytes that are inputtedfrom the RS frame encoder 112 into bits, which are then outputted to thesymbol encoder 402. The byte-bit converter 401 may also receivesignaling information including transmission parameters. The signalinginformation data bytes are also divided into bits so as to be outputtedto the symbol encoder 402. Herein, the signaling information includingtransmission parameters may be processed with the same data processingstep as that of the mobile service data. More specifically, thesignaling information may be inputted to the block processor 303 bypassing through the data randomizer 301 and the RS frame encoder 302.Alternatively, the signaling information may also be directly outputtedto the block processor 303 without passing though the data randomizer301 and the RS frame encoder 302.

The symbol encoder 402 corresponds to a G/H-rate encoder encoding theinputted data from G bits to H bits and outputting the data encoded atthe coding rate of G/H. According to the embodiment of the presentinvention, it is assumed that the symbol encoder 402 performs either acoding rate of 1/2 (also referred to as a 1/2-rate encoding process) oran encoding process at a coding rate of 1/4 (also referred to as a1/4-rate encoding process). The symbol encoder 402 performs one of1/2-rate encoding and 1/4-rate encoding on the inputted mobile servicedata and signaling information. Thereafter, the signaling information isalso recognized as the mobile service data and processed accordingly.

In case of performing the 1/2-rate coding process, the symbol encoder402 receives 1 bit and encodes the received 1 bit to 2 bits (i.e., 1symbol). Then, the symbol encoder 402 outputs the processed 2 bits (or 1symbol). On the other hand, in case of performing the 1/4-rate encodingprocess, the symbol encoder 402 receives 1 bit and encodes the received1 bit to 4 bits (i.e., 2 symbols). Then, the symbol encoder 402 outputsthe processed 4 bits (or 2 symbols).

FIG. 10 illustrates a detailed block diagram of the symbol encoder 402shown in FIG. 9. The symbol encoder 402 includes two delay units 501 and503 and three adders 502, 504, and 505. Herein, the symbol encoder 402encodes an input data bit U and outputs the coded bit U to 4 bits (u0 tou4). At this point, the data bit U is directly outputted as uppermostbit u0 and simultaneously encoded as lower bit u1u2u3 and thenoutputted. More specifically, the input data bit U is directly outputtedas the uppermost bit u0 and simultaneously outputted to the first andthird adders 502 and 505. The first adder 502 adds the input data bit Uand the output bit of the first delay unit 501 and, then, outputs theadded bit to the second delay unit 503. Then, the data bit delayed by apre-determined time (e.g., by 1 clock) in the second delay unit 503 isoutputted as lower bit u1 and simultaneously fed-back to the first delayunit 501. The first delay unit 501 delays the data bit fed-back from thesecond delay unit 503 by a pre-determined time (e.g., by 1 clock). Then,the first delay unit 501 outputs the delayed data bit to the first adder502 and the second adder 504. The second adder 504 adds the data bitsoutputted from the first and second delay units 501 and 503 as a lowerbit u2. The third adder 505 adds the input data bit U and the output ofthe second delay unit 503 and outputs the added data bit as a lower bitu3.

At this point, if the input data bit U corresponds to data encoded at a1/2-coding rate, the symbol encoder 402 configures a symbol with u1u0bits from the 4 output bits u0u1u2u3. Then, the symbol encoder 402outputs the newly configured symbol. Alternatively, if the input databit U corresponds to data encoded at a 1/4-coding rate, the symbolencoder 402 configures and outputs a symbol with bits u1u0 and, then,configures and outputs another symbol with bits u2u3. According toanother embodiment of the present invention, if the input data bit Ucorresponds to data encoded at a 1/4-coding rate, the symbol encoder 402may also configure and output a symbol with bits u1u0, and then repeatthe process once again and output the corresponding bits. According toyet another embodiment of the present invention, the symbol encoderoutputs all four output bits U u0u1u2u3. Then, when using the 1/2-codingrate, the symbol interleaver 403 located behind the symbol encoder 402selects only the symbol configured of bits u1u0 from the four outputbits u0u1u2u3. Alternatively, when using the 1/4-coding rate, the symbolinterleaver 403 may select the symbol configured of bits u1u0 and thenselect another symbol configured of bits u2u3. According to anotherembodiment, when using the 1/4-coding rate, the symbol interleaver 403may repeatedly select the symbol configured of bits u1u0.

The output of the symbol encoder 402 is inputted to the symbolinterleaver 403. Then, the symbol interleaver 403 performs blockinterleaving in symbol units on the data outputted from the symbolencoder 402. Any interleaver performing structural rearrangement (orrealignment) may be applied as the symbol interleaver 403 of the blockprocessor. However, in the present invention, a variable length symbolinterleaver that can be applied even when a plurality of lengths isprovided for the symbol, so that its order may be rearranged, may alsobe used.

FIG. 11 illustrates a symbol interleaver according to an embodiment ofthe present invention. Herein, the symbol interleaver according to theembodiment of the present invention corresponds to a variable lengthsymbol interleaver that may be applied even when a plurality of lengthsis provided for the symbol, so that its order may be rearranged.Particularly, FIG. 11 illustrates an example of the symbol interleaverwhen K=6 and L=8. Herein, K indicates a number of symbols that areoutputted for symbol interleaving from the symbol encoder 402. And, Lrepresents a number of symbols that are actually interleaved by thesymbol interleaver 403.

In the present invention, the symbol interleaver 403 should satisfy theconditions of L=2^(n) (wherein n is an integer) and of L≧K. If there isa difference in value between K and L, (L−K) number of null (or dummy)symbols is added, thereby creating an interleaving pattern. Therefore, Kbecomes a block size of the actual symbols that are inputted to thesymbol interleaver 403 in order to be interleaved. L becomes aninterleaving unit when the interleaving process is performed by aninterleaving pattern created from the symbol interleaver 403. Theexample of what is described above is illustrated in FIG. 11.

More specifically, FIG. 11( a) to FIG. 11( c) illustrate a variablelength interleaving process of a symbol interleaver shown in FIG. 9. Thenumber of symbols outputted from the symbol encoder 402 in order to beinterleaved is equal to 6 (i.e., K=6). In other words, 6 symbols areoutputted from the symbol encoder 402 in order to be interleaved. And,the actual interleaving unit (L) is equal to 8 symbols. Therefore, asshown in FIG. 11( a), 2 symbols are added to the null (or dummy) symbol,thereby creating the interleaving pattern. Equation 2 shown belowdescribed the process of sequentially receiving K number of symbols, theorder of which is to be rearranged, and obtaining an L value satisfyingthe conditions of L=2^(n) (wherein n is an integer) and of L≧K, therebycreating the interleaving so as to realign (or rearrange) the symbolorder.

In relation to all places, wherein 0≦i≦L−1,

P(i)={S×i×(i+1)/2} mod L  Equation 2

Herein, L≧K, L=2^(n), and n and S are integers. Referring to FIG. 11, itis assumed that S is equal to 89, and that L is equal to 8, and FIG. 11illustrates the created interleaving pattern and an example of theinterleaving process. As shown in FIG. 11( b), the order of K number ofinput symbols and (L−K) number of null symbols is rearranged by usingthe above-mentioned Equation 2. Then, as shown in FIG. 11( c), the nullbyte places are removed, so as to rearrange the order, by using Equation3 shown below. Thereafter, the symbol that is interleaved by therearranged order is then outputted to the symbol-byte converter.

if P(i)>K−1, then P(i)place is removed and rearranged  Equation 3

Subsequently, the symbol-byte converter 404 converts to bytes the mobileservice data symbols, having the rearranging of the symbol ordercompleted and then outputted in accordance with the rearranged order,and thereafter outputs the converted bytes to the group formatter 304.

FIG. 12A illustrates a block diagram showing the structure of a blockprocessor according to another embodiment of the present invention.Herein, the block processor includes an interleaving unit 610 and ablock formatter 620. The interleaving unit 610 may include a byte-symbolconverter 611, a symbol-byte converter 612, a symbol interleaver 613,and a symbol-byte converter 614. Herein, the symbol interleaver 613 mayalso be referred to as a block interleaver.

The byte-symbol converter 611 of the interleaving unit 610 converts themobile service data X outputted in byte units from the RS frame encoder302 to symbol units. Then, the byte-symbol converter 611 outputs theconverted mobile service data symbols to the symbol-byte converter 612and the symbol interleaver 613. More specifically, the byte-symbolconverter 611 converts each 2 bits of the inputted mobile service databyte (=8 bits) to 1 symbol and outputs the converted symbols. This isbecause the input data of the trellis encoding module 256 consist ofsymbol units configured of 2 bits. The relationship between the blockprocessor 303 and the trellis encoding module 256 will be described indetail in a later process. At this point, the byte-symbol converter 611may also receive signaling information including transmissionparameters. Furthermore, the signaling information bytes may also bedivided into symbol units and then outputted to the symbol-byteconverter 612 and the symbol interleaver 613.

The symbol-byte converter 612 groups 4 symbols outputted from thebyte-symbol converter 611 so as to configure a byte. Thereafter, theconverted data bytes are outputted to the block formatter 620. Herein,each of the symbol-byte converter 612 and the byte-symbol converter 611respectively performs an inverse process on one another. Therefore, theyield of these two blocks is offset. Accordingly, as shown in FIG. 12B,the input data X bypass the byte-symbol converter 611 and thesymbol-byte converter 612 and are directly inputted to the blockformatter 620. More specifically, the interleaving unit 610 of FIG. 12Bhas a structure equivalent to that of the interleaving unit shown inFIG. 12A. Therefore, the same reference numerals will be used in FIG.12A and FIG. 12B.

The symbol interleaver 613 performs block interleaving in symbol unitson the data that are outputted from the byte-symbol converter 611.Subsequently, the symbol interleaver 613 outputs the interleaved data tothe symbol-byte converter 614. Herein, any type of interleaver that canrearrange the structural order may be used as the symbol interleaver 613of the present invention. In the example given in the present invention,a variable length interleaver that may be applied for symbols having awide range of lengths, the order of which is to be rearranged. Forexample, the symbol interleaver of FIG. 11 may also be used in the blockprocessor shown in FIG. 12A and FIG. 12B.

The symbol-byte converter 614 outputs the symbols having the rearrangingof the symbol order completed, in accordance with the rearranged order.Thereafter, the symbols are grouped to be configured in byte units,which are then outputted to the block formatter 620. More specifically,the symbol-byte converter 614 groups 4 symbols outputted from the symbolinterleaver 613 so as to configure a data byte. As shown in FIG. 13, theblock formatter 620 performs the process of aligning the output of eachsymbol-byte converter 612 and 614 within the block in accordance with aset standard. Herein, the block formatter 620 operates in associationwith the trellis encoding module 256.

More specifically, the block formatter 620 decides the output order ofthe mobile service data outputted from each symbol-byte converter 612and 614 while taking into consideration the place (or order) of the dataexcluding the mobile service data that are being inputted, wherein themobile service data include main service data, known data, RS paritydata, and MPEG header data.

According to the embodiment of the present invention, the trellisencoding module 256 is provided with 12 trellis encoders. FIG. 14illustrates a block diagram showing the trellis encoding module 256according to the present invention. In the example shown in FIG. 14, 12identical trellis encoders are combined to the interleaver in order todisperse noise. Herein, each trellis encoder may be provided with apre-coder.

FIG. 15A illustrates the block processor 303 being concatenated with thetrellis encoding module 256. In the transmitting system, a plurality ofblocks actually exists between the pre-processor 230 including the blockprocessor 303 and the trellis encoding module 256, as shown in FIG. 3.Conversely, the receiving system considers the pre-processor 230 to beconcatenated with the trellis encoding module 256, thereby performingthe decoding process accordingly. However, the data excluding the mobileservice data that are being inputted to the trellis encoding module 256,wherein the mobile service data include main service data, known data,RS parity data, and MPEG header data, correspond to data that are addedto the blocks existing between the block processor 303 and the trellisencoding module 256. FIG. 15B illustrates an example of a data processor650 being positioned between the block processor 303 and the trellisencoding module 256, while taking the above-described instance intoconsideration.

Herein, when the interleaving unit 610 of the block processor 303performs a 1/2-rate encoding process, the interleaving unit 610 may beconfigured as shown in FIG. 12A (or FIG. 12B). Referring to FIG. 3, forexample, the data processor 650 may include a group formatter 304, adata deinterleaver 305, a packet formatter 306, a packet multiplexer240, and a post-processor 250, wherein the post-processor 250 includes adata randomizer 251, a RS encoder/non-systematic RS encoder 252, a datainterleaver 253, a parity replacer 254, and a non-systematic RS encoder255.

At this point, the trellis encoding module 256 symbolizes the data thatare being inputted so as to divide the symbolized data and to send thedivided data to each trellis encoder in accordance with a pre-definedmethod. Herein, one byte is converted into 4 symbols, each beingconfigured of 2 bits. Also, the symbols created from the single databyte are all transmitted to the same trellis encoder. Accordingly, eachtrellis encoder pre-codes an upper bit of the input symbol, which isthen outputted as the uppermost output bit C2. Alternatively, eachtrellis encoder trellis-encodes a lower bit of the input symbol, whichis then outputted as two output bits C1 and C0. The block formatter 620is controlled so that the data byte outputted from each symbol-byteconverter can be transmitted to different trellis encoders.

Hereinafter, the operation of the block formatter 620 will now bedescribed in detail with reference to FIG. 9 to FIG. 12. Referring toFIG. 12A, for example, the data byte outputted from the symbol-byteconverter 612 and the data byte outputted from the symbol-byte converter614 are inputted to different trellis encoders of the trellis encodingmodule 256 in accordance with the control of the block formatter 620.Hereinafter, the data byte outputted from the symbol-byte converter 612will be referred to as X, and the data byte outputted from thesymbol-byte converter 614 will be referred to as Y, for simplicity.Referring to FIG. 13( a), each number (i.e., 0 to 11) indicates thefirst to twelfth trellis encoders of the trellis encoding module 256,respectively.

In addition, the output order of both symbol-byte converters arearranged (or aligned) so that the data bytes outputted from thesymbol-byte converter 612 are respectively inputted to the 0^(th) to5^(th) trellis encoders (0 to 5) of the trellis encoding module 256, andthat the data bytes outputted from the symbol-byte converter 614 arerespectively inputted to the 6^(th) to 11^(th) trellis encoders (6 to11) of the trellis encoding module 256. Herein, the trellis encodershaving the data bytes outputted from the symbol-byte converter 612allocated therein, and the trellis encoders having the data bytesoutputted from the symbol-byte converter 614 allocated therein aremerely examples given to simplify the understanding of the presentinvention. Furthermore, according to an embodiment of the presentinvention, and assuming that the input data of the block processor 303correspond to a block configured of 12 bytes, the symbol-byte converter612 outputs 12 data bytes from X0 to X11, and the symbol-byte converter614 outputs 12 data bytes from Y0 to Y11.

FIG. 13( b) illustrates an example of data being inputted to the trellisencoding module 256. Particularly, FIG. 13( b) illustrates an example ofnot only the mobile service data but also the main service data and RSparity data being inputted to the trellis encoding module 256, so as tobe distributed to each trellis encoder. More specifically, the mobileservice data outputted from the block processor 303 pass through thegroup formatter 304, from which the mobile service data are mixed withthe main service data and RS parity data and then outputted, as shown inFIG. 13( a). Accordingly, each data byte is respectively inputted to the12 trellis encoders in accordance with the positions (or places) withinthe data group after being data-interleaved.

Herein, when the output data bytes X and Y of the symbol-byte converters612 and 614 are allocated to each respective trellis encoder, the inputof each trellis encoder may be configured as shown in FIG. 13( b). Morespecifically, referring to FIG. 13( b), the six mobile service databytes (X0 to X5) outputted from the symbol-byte converter 612 aresequentially allocated (or distributed) to the first to sixth trellisencoders (0 to 5) of the trellis encoding module 256. Also, the 2 mobileservice data bytes Y0 and Y1 outputted from the symbol-byte converter614 are sequentially allocated to the 7^(th) and 8^(th) trellis encoders(6 and 7) of the trellis encoding module 256. Thereafter, among the 5main service data bytes, 4 data bytes are sequentially allocated to the9^(th) and 12^(th) trellis encoders (8 to 11) of the trellis encodingmodule 256. Finally, the remaining 1 byte of the main service data byteis allocated once again to the first trellis encoder (0).

It is assumed that the mobile service data, the main service data, andthe RS parity data are allocated to each trellis encoder, as shown inFIG. 13( b). It is also assumed that, as described above, the input ofthe block processor 303 is configured of 12 bytes, and that 12 bytesfrom X0 to X11 are outputted from the symbol-byte converter 612, andthat 12 bytes from Y0 to Y11 are outputted from the symbol-byteconverter 614. In this case, as shown in FIG. 13( c), the blockformatter 620 arranges the data bytes that are to be outputted from thesymbol-byte converters 612 and 614 by the order of X0 to X5, Y0, Y1, X6to X10, Y2 to Y7, X11, and Y8 to Y11. More specifically, the trellisencoder that is to perform the encoding process is decided based uponthe position (or place) within the transmission frame in which each databyte is inserted. At this point, not only the mobile service data butalso the main service data, the MPEG header data, and the RS parity dataare also inputted to the trellis encoding module 256. Herein, it isassumed that, in order to perform the above-described operation, theblock formatter 620 is informed of (or knows) the information on thedata group format after the data-interleaving process.

FIG. 16 illustrates a block diagram of the block processor performing anencoding process at a coding rate of 1/N according to an embodiment ofthe present invention. Herein, the block processor includes (N−1) numberof symbol interleavers 741 to 74N−1, which are configured in a parallelstructure. More specifically, the block processor having the coding rateof 1/N consists of a total of N number of branches (or paths) includinga branch (or path), which is directly transmitted to the block formatter730. In addition, the symbol interleaver 741 to 74N−1 of each branch mayeach be configured of a different symbol interleaver. Furthermore, (N−1)number of symbol-byte converter 751 to 75N−1 each corresponding to each(N−1) number of symbol interleavers 741 to 74N−1 may be included at theend of each symbol interleaver, respectively. Herein, the output data ofthe (N−1) number of symbol-byte converter 751 to 75N−1 are also inputtedto the block formatter 730.

In the example of the present invention, N is equal to or smaller than12. If N is equal to 12, the block formatter 730 may align the outputdata so that the output byte of the 12^(th) symbol-byte converter 75N−1is inputted to the 12^(th) trellis encoder. Alternatively, if N is equalto 3, the block formatter 730 may arranged the output order, so that thedata bytes outputted from the symbol-byte converter 720 are inputted tothe 1^(st) to 4^(th) trellis encoders of the trellis encoding module256, and that the data bytes outputted from the symbol-byte converter751 are inputted to the 5^(th) to 8^(th) trellis encoders, and that thedata bytes outputted from the symbol-byte converter 752 are inputted tothe 9^(th) to 12^(th) trellis encoders. At this point, the order of thedata bytes outputted from each symbol-byte converter may vary inaccordance with the position within the data group of the data otherthan the mobile service data, which are mixed with the mobile servicedata that are outputted from each symbol-byte converter.

FIG. 17 illustrates a detailed block diagram showing the structure of ablock processor according to another embodiment of the presentinvention. Herein, the block formatter is removed from the blockprocessor so that the operation of the block formatter may be performedby a group formatter. More specifically, the block processor of FIG. 17may include a byte-symbol converter 810, symbol-byte converters 820 and840, and a symbol interleaver 830. In this case, the output of eachsymbol-byte converter 820 and 840 is inputted to the group formatter850.

Also, the block processor may obtain a desired coding rate by addingsymbol interleavers and symbol-byte converters. If the system designerwishes a coding rate of 1/N, the block processor needs to be providedwith a total of N number of branches (or paths) including a branch (orpath), which is directly transmitted to the block formatter 850, and(N−1) number of symbol interleavers and symbol-byte convertersconfigured in a parallel structure with (N−1) number of branches. Atthis point, the group formatter 850 inserts place holders ensuring thepositions (or places) for the MPEG header, the non-systematic RS parity,and the main service data. And, at the same time, the group formatter850 positions the data bytes outputted from each branch of the blockprocessor.

The number of trellis encoders, the number of symbol-byte converters,and the number of symbol interleavers proposed in the present inventionare merely exemplary. And, therefore, the corresponding numbers do notlimit the spirit or scope of the present invention. It is apparent tothose skilled in the art that the type and position of each data bytebeing allocated to each trellis encoder of the trellis encoding module256 may vary in accordance with the data group format. Therefore, thepresent invention should not be understood merely by the examples givenin the description set forth herein. The mobile service data that areencoded at a coding rate of 1/N and outputted from the block processor303 are inputted to the group formatter 304. Herein, in the example ofthe present invention, the order of the output data outputted from theblock formatter of the block processor 303 are aligned and outputted inaccordance with the position of the data bytes within the data group.

Signaling Information Processing

The transmitter 200 according to the present invention may inserttransmission parameters by using a plurality of methods and in aplurality of positions (or places), which are then transmitted to thereceiving system. For simplicity, the definition of a transmissionparameter that is to be transmitted from the transmitter to thereceiving system will now be described. The transmission parameterincludes data group information, region information within a data group,the number of RS frames configuring a super frame (i.e., a super framesize (SFS)), the number of RS parity data bytes (P) for each columnwithin the RS frame, whether or not a checksum, which is added todetermine the presence of an error in a row direction within the RSframe, has been used, the type and size of the checksum if the checksumis used (presently, 2 bytes are added to the CRC), the number of datagroups configuring one RS frame—since the RS frame is transmitted to oneburst section, the number of data groups configuring the one RS frame isidentical to the number of data groups within one burst (i.e., burstsize (BS)), a turbo code mode, and a RS code mode.

Also, the transmission parameter required for receiving a burst includesa burst period—herein, one burst period corresponds to a value obtainedby counting the number of fields starting from the beginning of acurrent burst until the beginning of a next burst, a positioning orderof the RS frames that are currently being transmitted within a superframe (i.e., a permuted frame index (PFI)) or a positioning order ofgroups that are currently being transmitted within a RS frame (burst)(i.e., a group index (GI)), and a burst size. Depending upon the methodof managing a burst, the transmission parameter also includes the numberof fields remaining until the beginning of the next burst (i.e., time tonext burst (TNB)). And, by transmitting such information as thetransmission parameter, each data group being transmitted to thereceiving system may indicate a relative distance (or number of fields)between a current position and the beginning of a next burst.

The information included in the transmission parameter corresponds toexamples given to facilitate the understanding of the present invention.Therefore, the proposed examples do not limit the scope or spirit of thepresent invention and may be easily varied or modified by anyone skilledin the art. According to the first embodiment of the present invention,the transmission parameter may be inserted by allocating a predeterminedregion of the mobile service data packet or the data group. In thiscase, the receiving system performs synchronization and equalization ona received signal, which is then decoded by symbol units. Thereafter,the packet deformatter may separate the mobile service data and thetransmission parameter so as to detect the transmission parameter.According to the first embodiment, the transmission parameter may beinserted from the group formatter 304 and then transmitted.

According to the second embodiment of the present invention, thetransmission parameter may be multiplexed with another type of data. Forexample, when known data are multiplexed with the mobile service data, atransmission parameter may be inserted, instead of the known data, in aplace (or position) where a known data byte is to be inserted.Alternatively, the transmission parameter may be mixed with the knowndata and then inserted in the place where the known data byte is to beinserted. According to the second embodiment, the transmission parametermay be inserted from the group formatter 304 or from the packetformatter 306 and then transmitted.

According to a third embodiment of the present invention, thetransmission parameter may be inserted by allocating a portion of areserved region within a field synchronization segment of a transmissionframe. In this case, since the receiving system may perform decoding ona receiving signal by symbol units before detecting the transmissionparameter, the transmission parameter having information on theprocessing methods of the block processor 303 and the group formatter304 may be inserted in a reserved field of a field synchronizationsignal. More specifically, the receiving system obtains fieldsynchronization by using a field synchronization segment so as to detectthe transmission parameter from a pre-decided position. According to thethird embodiment, the transmission parameter may be inserted from thesynchronization multiplexer 240 and then transmitted.

According to the fourth embodiment of the present invention, thetransmission parameter may be inserted in a layer (or hierarchicalregion) higher than a transport stream (TS) packet. In this case, thereceiving system should be able to receive a signal and process thereceived signal to a layer higher than the TS packet in advance. At thispoint, the transmission parameter may be used to certify thetransmission parameter of a currently received signal and to provide thetransmission parameter of a signal that is to be received in a laterprocess.

In the present invention, the variety of transmission parametersassociated with the transmission signal may be inserted and transmittedby using the above-described methods according to the first to fourthembodiment of the present invention. At this point, the transmissionparameter may be inserted and transmitted by using only one of the fourembodiments described above, or by using a selection of theabove-described embodiments, or by using all of the above-describedembodiments. Furthermore, the information included in the transmissionparameter may be duplicated and inserted in each embodiment.Alternatively, only the required information may be inserted in thecorresponding position of the corresponding embodiment and thentransmitted. Furthermore, in order to ensure robustness of thetransmission parameter, a block encoding process of a short cycle (orperiod) may be performed on the transmission parameter and, then,inserted in a corresponding region. The method for performing ashort-period block encoding process on the transmission parameter mayinclude, for example, Kerdock encoding, BCH encoding, RS encoding, andrepetition encoding of the transmission parameter. Also, a combinationof a plurality of block encoding methods may also be performed on thetransmission parameter.

The transmission parameters may be grouped to create a block code of asmall size, so as to be inserted in a byte place allocated within thedata group for signaling and then transmitted. However, in this case,the block code passes through the block decoded from the receiving endso as to obtain a transmission parameter value. Therefore, thetransmission parameters of the turbo code mode and the RS code mode,which are required for block decoding, should first be obtained.Accordingly, the transmission parameters associated with a particularmode may be inserted in a specific section of a known data region. And,in this case, a correlation of with a symbol may be used for a fasterdecoding process. The receiving system refers to the correlation betweeneach sequence and the currently received sequences, thereby determiningthe encoding mode and the combination mode.

Meanwhile, when the transmission parameter is inserted in the fieldsynchronization segment region or the known data region and thentransmitted, and when the transmission parameter has passed through thetransmission channel, the reliability of the transmission parameter isdeteriorated. Therefore, one of a plurality of pre-defined patterns mayalso be inserted in accordance with the corresponding transmissionparameter. Herein, the receiving system performs a correlationcalculation between the received signal and the pre-defined patterns soas to recognize the transmission parameter. For example, it is assumedthat a burst including 5 data groups is pre-decided as pattern A basedupon an agreement between the transmitting system and the receivingsystem. In this case, the transmitting system inserts and transmitspattern A, when the number of groups within the burst is equal to 5.Thereafter, the receiving system calculates a correlation between thereceived data and a plurality of reference patterns including pattern A,which was created in advance. At this point, if the correlation valuebetween the received data and pattern A is the greatest, the receiveddata indicates the corresponding parameter, and most particularly, thenumber of groups within the burst. At this point, the number of groupsmay be acknowledged as 5. Hereinafter, the process of inserting andtransmitting the transmission parameter will now be described accordingto first, second, and third embodiments of the present invention.

First Embodiment

FIG. 18 illustrates a schematic diagram of the group formatter 304receiving the transmission parameter and inserting the receivedtransmission parameter in region A of the data group according to thepresent invention. Herein, the group formatter 304 receives mobileservice data from the block processor 303. Conversely, the transmissionparameter is processed with at least one of a data randomizing process,a RS frame encoding process, and a block processing process, and maythen be inputted to the group formatter 304. Alternatively, thetransmission parameter may be directly inputted to the group formatter304 without being processed with any of the above-mentioned processes.In addition, the transmission parameter may be provided from the servicemultiplexer 100. Alternatively, the transmission parameter may also begenerated and provided from within the transmitter 200. The transmissionparameter may also include information required by the receiving systemin order to receive and process the data included in the data group. Forexample, the transmission parameter may include data group information,and multiplexing information.

The group formatter 304 inserts the mobile service data and transmissionparameter which are to be inputted to corresponding regions within thedata group in accordance with a rule for configuring a data group. Forexample, the transmission parameter passes through a block encodingprocess of a short period and is, then, inserted in region A of the datagroup. Particularly, the transmission parameter may be inserted in apre-arranged and arbitrary position (or place) within region A. If it isassumed that the transmission parameter has been block encoded by theblock processor 303, the block processor 303 performs the same dataprocessing operation as the mobile service data, more specifically,either a 1/2-rate encoding or 1/4-rate encoding process on the signalinginformation including the transmission parameter. Thereafter, the blockprocessor 303 outputs the processed transmission parameter to the groupformatter 304. Thereafter, the signaling information is also recognizedas the mobile service data and processed accordingly.

FIG. 19 illustrates a block diagram showing an example of the blockprocessor receiving the transmission parameter and processing thereceived transmission parameter with the same process as the mobileservice data. Particularly, FIG. 19 illustrates an example showing thestructure of FIG. 9 further including a signaling information provider411 and multiplexer 412. More specifically, the signaling informationprovider 411 outputs the signaling information including thetransmission parameter to the multiplexer 412. The multiplexer 412multiplexes the signaling information and the output of the RS frameencoder 302. Then, the multiplexer 412 outputs the multiplexed data tothe byte-bit converter 401.

The byte-bit converter 401 divides the mobile service data bytes orsignaling information byte outputted from the multiplexer 412 into bits,which are then outputted to the symbol encoder 402. The subsequentoperations are identical to those described in FIG. 9. Therefore, adetailed description of the same will be omitted for simplicity. If anyof the detailed structures of the block processor 303 shown in FIG. 12,FIG. 15, FIG. 16, and FIG. 17, the signaling information provider 411and the multiplexer 412 may be provided behind the byte-symbolconverter.

Second Embodiment

Meanwhile, when known data generated from the group formatter inaccordance with a pre-decided rule are inserted in a correspondingregion within the data group, a transmission parameter may be insertedin at least a portion of a region, where known data may be inserted,instead of the known data. For example, when a long known data sequenceis inserted at the beginning of region A within the data group, atransmission parameter may be inserted in at least a portion of thebeginning of region A instead of the known data. A portion of the knowndata sequence that is inserted in the remaining portion of region A,excluding the portion in which the transmission parameter is inserted,may be used to detect a starting point of the data group by thereceiving system. Alternatively, another portion of region A may be usedfor channel equalization by the receiving system.

In addition, when the transmission parameter is inserted in the knowndata region instead of the actual known data. The transmission parametermay be block encoded in short periods and then inserted. Also, asdescribed above, the transmission parameter may also be inserted basedupon a pre-defined pattern in accordance with the transmissionparameter. If the group formatter 304 inserts known data place holdersin a region within the data group, wherein known data may be inserted,instead of the actual known data, the transmission parameter may beinserted by the packet formatter 306. More specifically, when the groupformatter 304 inserts the known data place holders, the packet formatter306 may insert the known data instead of the known data place holders.Alternatively, when the group formatter 304 inserts the known data, theknown data may be directly outputted without modification.

FIG. 20 illustrates a block diagram showing the structure of a packetformatter 306 being expanded so that the packet formatter 306 can insertthe transmission parameter according to an embodiment of the presentinvention. More specifically, the structure of the packet formatter 306further includes a known data generator 351 and a signaling multiplexer352. Herein, the transmission parameter that is inputted to thesignaling multiplexer 352 may include information on the length of acurrent burst, information indicating a starting point of a next burst,positions in which the groups within the burst exist and the lengths ofthe groups, information on the time from the current group and the nextgroup within the burst, and information on known data.

The signaling multiplexer 352 selects one of the transmission parameterand the known data generated from the known data generator 351 and,then, outputs the selected data to the packet formatter 306. The packetformatter 306 inserts the known data or transmission parameter outputtedfrom the signaling multiplexer 352 into the known data place holdersoutputted from the data interleaver 305. Then, the packet formatter 306outputs the processed data. More specifically, the packet formatter 306inserts a transmission parameter in at least a portion of the known dataregion instead of the known data, which is then outputted. For example,when a known data place holder is inserted at a beginning portion ofregion A within the data group, a transmission parameter may be insertedin a portion of the known data place holder instead of the actual knowndata.

Also, when the transmission parameter is inserted in the known dataplace holder instead of the known data, the transmission parameter maybe block encoded in short periods and inserted. Alternatively, apre-defined pattern may be inserted in accordance with the transmissionparameter. More specifically, the signaling multiplexer 352 multiplexesthe known data and the transmission parameter (or the pattern defined bythe transmission parameter) so as to configure a new known datasequence. Then, the signaling multiplexer 352 outputs the newlyconfigured known data sequence to the packet formatter 306. The packetformatter 306 deletes the main service data place holder and RS parityplace holder from the output of the data interleaver 305, and creates amobile service data packet of 188 bytes by using the mobile servicedata, MPEG header, and the output of the signaling multiplexer. Then,the packet formatter 306 outputs the newly created mobile service datapacket to the packet multiplexer 240.

In this case, the region A of each data group has a different known datapattern. Therefore, the receiving system separates only the symbol in apre-arranged section of the known data sequence and recognizes theseparated symbol as the transmission parameter. Herein, depending uponthe design of the transmitting system, the known data may be inserted indifferent blocks, such as the packet formatter 306, the group formatter304, or the block processor 303. Therefore, a transmission parameter maybe inserted instead of the known data in the block wherein the knowndata are to be inserted.

According to the second embodiment of the present invention, atransmission parameter including information on the processing method ofthe block processor 303 may be inserted in a portion of the known dataregion and then transmitted. In this case, a symbol processing methodand position of the symbol for the actual transmission parameter symbolare already decided. Also, the position of the transmission parametersymbol should be positioned so as to be transmitted or received earlierthan any other data symbols that are to be decoded. Accordingly, thereceiving system may detect the transmission symbol before the datasymbol decoding process, so as to use the detected transmission symbolfor the decoding process.

Third Embodiment

Meanwhile, the transmission parameter may also be inserted in the fieldsynchronization segment region and then transmitted. FIG. 21 illustratesa block diagram showing the synchronization multiplexer being expandedin order to allow the transmission parameter to be inserted in the fieldsynchronization segment region. Herein, a signaling multiplexer 261 isfurther included in the synchronization multiplexer 260. Thetransmission parameter of the general VSB method is configured of 2fields. More specifically, each field is configured of one fieldsynchronization segment and 312 data segments. Herein, the first 4symbols of a data segment correspond to the segment synchronizationportion, and the first data segment of each field corresponds to thefield synchronization portion.

One field synchronization signal is configured to have the length of onedata segment. The data segment synchronization pattern exists in thefirst 4 symbols, which are then followed by pseudo random sequences PN511, PN 63, PN 63, and PN 63. The next 24 symbols include informationassociated with the VSB mode. Additionally, the 24 symbols that includeinformation associated with the VSB mode are followed by the remaining104 symbols, which are reserved symbols. Herein, the last 12 symbols ofa previous segment are copied and positioned as the last 12 symbols inthe reserved region. In other words, only the 92 symbols in the fieldsynchronization segment are the symbols that correspond to the actualreserved region.

Therefore, the signaling multiplexer 261 multiplexes the transmissionparameter with an already-existing field synchronization segment symbol,so that the transmission parameter can be inserted in the reservedregion of the field synchronization segment. Then, the signalingmultiplexer 261 outputs the multiplexed transmission parameter to thesynchronization multiplexer 260. The synchronization multiplexer 260multiplexes the segment synchronization symbol, the data symbols, andthe new field synchronization segment outputted from the signalingmultiplexer 261, thereby configuring a new transmission frame. Thetransmission frame including the field synchronization segment, whereinthe transmission parameter is inserted, is outputted to the transmissionunit 270. At this point, the reserved region within the fieldsynchronization segment for inserting the transmission parameter maycorrespond to a portion of or the entire 92 symbols of the reservedregion. Herein, the transmission parameter being inserted in thereserved region may, for example, include information identifying thetransmission parameter as the main service data, the mobile servicedata, or a different type of mobile service data.

If the information on the processing method of the block processor 303is transmitted as a portion of the transmission parameter, and when thereceiving system wishes to perform a decoding process corresponding tothe block processor 303, the receiving system should be informed of suchinformation on the block processing method in order to perform thedecoding process. Therefore, the information on the processing method ofthe block processor 303 should already be known prior to the blockdecoding process. Accordingly, as described in the third embodiment ofthe present invention, when the transmission parameter having theinformation on the processing method of the block processor 303 (and/orthe group formatter 304) is inserted in the reserved region of the fieldsynchronization signal and then transmitted, the receiving system iscapable of detecting the transmission parameter prior to performing theblock decoding process on the received signal.

Receiving System

FIG. 22 illustrates a block diagram showing a structure of a digitalbroadcast receiving system according to the present invention. Thedigital broadcast receiving system of FIG. 22 uses known datainformation, which is inserted in the mobile service data section and,then, transmitted by the transmitting system, so as to perform carriersynchronization recovery, frame synchronization recovery, and channelequalization, thereby enhancing the receiving performance. Referring toFIG. 22, the digital broadcast receiving system may include a tuner 901,a demodulator 902, an equalizer 903, a known data detector 904, a blockdecoder 905, a data deformatter 906, a RS frame decoder 907, aderandomizer 908, a data deinterleaver 909, a RS decoder 910, and a dataderandomizer 911. Herein, for simplicity of the description of thepresent invention, the data deformatter 906, the RS frame decoder 907,and the derandomizer 908 will be collectively referred to as a mobileservice data processing unit. And, the data deinterleaver 909, the RSdecoder 910, and the data derandomizer 911 will be collectively referredto as a main service data processing unit.

More specifically, the tuner 901 tunes a frequency of a particularchannel and down-converts the tuned frequency to an intermediatefrequency (IF) signal. Then, the tuner 901 outputs the down-converted IFsignal to the demodulator 902 and the known data detector 904. Thedemodulator 902 performs self gain control, carrier recovery, and timingrecovery processes on the inputted IF signal, thereby modifying the IFsignal to a baseband signal. Then, the demodulator 902 outputs the newlycreated baseband signal to the equalizer 903 and the known data detector904. The equalizer 903 compensates the distortion of the channelincluded in the demodulated signal and then outputs theerror-compensated signal to the block decoder 905.

At this point, the known data detector 904 detects the known sequenceposition inserted by the transmitting end from the input/output data ofthe demodulator 902 (i.e., the data prior to the demodulation process orthe data after the demodulation process). Thereafter, the known datadetector 904 outputs the detected known sequence position indicatoralong with the known data generated from the detected known sequenceposition to the demodulator 902 and the equalizer 903. Herein, theequalizer 903 may also receive only the known sequence positionindicators having the same pattern. Also, the known data detector 904outputs a set of information to the block decoder 905. This set ofinformation is used to allow the block decoder 905 of the receivingsystem to identify the mobile service data that are processed withadditional encoding from the transmitting system and the main servicedata that are not processed with additional encoding.

In addition, although the connection status is not shown in FIG. 22, theinformation detected from the known data detector 904 may be usedthroughout the entire receiving system and may also be used in the datadeformatter 906 and the RS frame decoder 907. The demodulator 902 usesthe known sequence position indicator and the known data during thetiming and/or carrier recovery, thereby enhancing the demodulatingperformance. Similarly, the equalizer 903 uses the known sequenceposition indicator and the known data so as to enhance the equalizingperformance. Moreover, the decoding result of the block decoder 905 maybe fed-back to the equalizer 903, thereby enhancing the equalizingperformance. The equalizer 903 may perform channel equalization by usinga plurality of methods. In the description of the present invention, thechannel equalizing process will be described according to a plurality ofembodiments. At this point, the transmitting system may periodicallyinsert and transmit known data within a transmission frame, as shown inFIG. 6A.

FIG. 23 illustrates an example of known data sequence being periodicallyinserted and transmitted in-between actual data by the transmittingsystem. Referring to FIG. 23, D represents the number of known datasymbols, and E represents the number of general data symbols. Therefore,D number of known data symbols are inserted and transmitted at a periodof (D+E) symbols. Herein, E may correspond to data that are not knowndata, such as mobile service data, main service data, or a combinationof mobile service data and main service data. In order to bedifferentiated from the known data, data corresponding to E willhereinafter be referred to as “general data”.

Referring to FIG. 23, known data sequence having the same pattern areincluded in each known data section that is being periodically inserted.Herein, the length of the known data sequence having identical datapatterns may be either equal to or different from the length of theentire (or total) known data sequence of the corresponding known datasection (or block). If the two lengths are different from one another,the length of the entire known data sequence is longer than the lengthof the known data sequence having identical data patterns. In this case,the same known data sequences are included in the entire known datasequence. For example, if a known data pattern F is inserted for channelequalizing, each known data section includes at least one known datapattern F.

Accordingly, when the known data are regularly inserted in the generaldata as described above, the channel equalizer included in the digitalbroadcast receiving system may use the inserted known data as trainingsequences, which may be used either for an accurate decision value orfor estimating an impulse response of a channel. Meanwhile, when theknown data having the same patterns are regularly inserted, each knowndata section (or block or interval) may be used as a guard interval inthe channel equalizer according to the present invention. The guardinterval prevents interference that occurs between blocks due to amultiple path channel. This is because the known data of the known datasection located behind the data block corresponding to the (D+E) symbolmay be considered as being copied in front of the data blockcorresponding to the (D+E) symbol, as shown in FIG. 23.

The above-described structure is referred to as a cyclic prefix. Thisstructure provides circular convolution in a time domain between a datablock transmitted from the transmitting system and a channel impulseresponse. Accordingly, this facilitates the channel equalizer of thereceiving system to perform channel equalization in a frequency domainby using a fast fourier transform (FFT) and an inverse fast fouriertransform (IFFT). More specifically, when viewed in the frequencydomain, the data block received by the receiving system is expressed asa multiplication of the data block and the channel impulse response.Therefore, when performing the channel equalization, by multiplying theinverse of the channel in the frequency domain, the channel equalizationmay be performed more easily.

FIG. 24 and FIG. 25 illustrate examples of the known data section actingas a guard section in a multiple path channel environment, therebypreventing interference in adjacent (or neighboring) blocks. Referringto FIG. 24 and FIG. 25, the path having the largest amount of energywill be referred to as a main path for simplicity. Herein, as shown inFIG. 24( a), a ghost being received after the main path is defined as apost-ghost. Alternatively, as shown in FIG. 25( a), a ghost beingreceived before the main path is defined as a pre-ghost. Referring tothe drawings, assuming that both the post-ghost and the pre-ghost exist,the FFT block which corresponds to a section performing FFT operationsis set to have a guard section located both in front of and behind thegeneral data. Herein, the known data located in front of the generaldata within the FFT block is referred to as D2, and the known datalocated behind the general data within the FFT block is referred to asD1.

FIG. 24 illustrates correlations between a main signal and a ghostsignal when a post-ghost exists. More specifically, FIG. 24( b)illustrates an example of a main signal, and FIG. 24( c) illustrates anexample of the post-ghost signal. Referring to FIG. 24, when a timedifference between the main signal and a ghost is t0, and when t0 issmaller than D2, the known data pattern that is regularly transmitted isidentical. Therefore, signal B which is added to the main signal withinthe FFT block by the ghost corresponds to the same data as the endportion A of the main signal. More specifically, since a signal having acircularly shifted data block is added to the current data block of themain signal within the FFT block, interference from an adjacent datablock may be prevented. In other words, the effect (or influence) of theghost is shown in a circular convolution form within the time domain ofthe main signal and an impulse response of a channel.

FIG. 25 illustrates correlations between a main signal and a ghostsignal when a pre-ghost exists. More specifically, FIG. 25( b)illustrates an example of a main signal, and FIG. 25( c) illustrates anexample of the pre-ghost signal. Referring to FIG. 25, which is similarto the example of the post-ghost shown in FIG. 24, when the timedifference t0 between the main signal and the ghost is smaller than D1,the effect (or influence) of the ghost is shown in a circularconvolution form of the main signal and the impulse response of achannel.

As described above, when the time difference between the main signal andthe pre-ghost is smaller than the D1 symbol time, or when the timedifference between the main signal and the post-ghost is smaller thanthe D2 symbol time, the effect of the ghost is shown within the FFTblock in a circular convolution form. More specifically, when a delayspread of a multiple path (or ghost) is smaller than the guard section,the effect of the multiple path becomes a circular convolution form ofthe general data block (or FFT block) and the channel impulse responsewithin the time domain. Accordingly, in the frequency domain, the effectof the multiple path is shown as a multiplication of the data block andthe channel impulse response. Therefore, by using such characteristicschannel equalization may be easily performed in the frequency domain.Thus, the structure of the channel equalization may be simplified. Thepresent invention may uses an indirect channel equalizing method,wherein known data sequences allocated in identical patterns during apre-determined cycle period are used to estimate a channel impulseresponse (CIR). Herein, an indirect channel equalizing method refers toa method of performing channel equalization by estimating the impulseresponse of a channel, thereby obtaining an equalization coefficient.Alternatively, a direct channel equalizing method refers to a method ofperforming channel equalization by obtaining an error from a channelequalized signal, thereby updating the equalization coefficient.

FIG. 26 illustrates a block diagram of an example of a channel equalizerusing an indirect channel equalizing method according to the presentinvention. Herein, the channel equalizing process is performed in afrequency domain. Referring to FIG. 26, the channel equalizer includes afirst fast fourier transform (FFT) unit 1001, a CIR estimator 1002, asecond FFT unit 1003, a coefficient calculator 1004, a distortioncompensator 1005, and an inverse fast fourier transform (IFFT) unit1006. More specifically, when a series of identical known data sequencesis regularly inserted in the general data sequence and then transmitted,the first FFT unit 1001 performs FFT by FFT block units on the receiveddata. Then, the first FFT unit 501 converts the processed data tofrequency domain data, which are then outputted to the distortioncompensator 1005. Herein, assuming that both post-ghost and pre-ghostexist, the FFT block section is set to have part of the known datacorresponding to the known data section positioned both before (or infront of) and after (or behind) the general data. Any device performingmultiplication of complex numbers may be used as the distortioncompensator 1005.

The CIR estimator 1002 uses the data received during the known datasection and the known data, which are known by the receiver according toan agreement between the receiving system and the transmitting system,in order to estimate a channel. Then, the CIR estimator 1002 outputs theestimated channel to the second FFT unit 1003. For this, the CIRestimator 1002 receives known sequence position indicators from theknown data detector 904. In this case, the CIR estimator 1002 may alsoreceive known sequence position indicators having identical patterns.Furthermore, since the known data sequence of the known data section (orblock) corresponds to data already known according to an agreementbetween the receiving system and the transmitting system, the known datasequence may be stored in advance (or pre-stored) in the CIR estimator1002. Alternatively, the known data sequences generated from the knowndata detector 904 may also be received.

According to the embodiment of the present invention, the known datasequences that is to be inserted as a reference data sequence is storedin advance in the CIR estimator 1002. Also, in addition to theembodiment shown in FIG. 26, the example of storing the known data inthe CIR estimator 1002 in advance, or receiving the known data from theknown data detector 904 may also be applied in other CIR estimatorsaccording to other embodiments of the present invention. Morespecifically, the CIR estimator 1002 performs the channel estimationprocess by using the known data being received only during the knowndata section and the pre-stored known data sequence. In other words, theCIR estimator 1002 estimates an impulse response of a discreteequalization channel through which a signal transmitted from thetransmitter is assumed to have passed during the known data section.Thereafter, the CIR estimator 1002 outputs the estimated result to thesecond FFT unit 1003.

The second FFT unit 1003 performs a FFT process on the estimated channeland converts the processed channel to the frequency domain. Thereafter,the processed channel is outputted to the coefficient calculator 1004.The coefficient calculator 1004 uses the channel impulse response of thefrequency domain to calculate the equalizer coefficient and outputs thecalculated result to the distortion compensator 1005. The distortioncompensator 1005 performs complex multiplication on the output of thecoefficient calculator 1004 and the output of the first FFT unit 1001after each frequency bin, thereby compensating the channel distortion ofthe data outputted from the first FFT unit 1001. Then, the distortioncompensator 1005 outputs the processed data to the IFFT unit 1006. TheIFFT unit 1006 performs an IFFT process of the data having the channeldistortion compensated by the distortion compensator 1005, therebyconverting the multiplied result to a time domain signal.

The coefficient calculator 1004 of FIG. 26 may calculate the equalizercoefficient by using a zero forcing method, which simply calculates theinverse of a channel. Alternatively, the coefficient calculator 1004 mayalso calculate the equalizer coefficient by using a minimum mean squarederror (MMSE) method, which estimates the amount of noise in the channelso as to minimize a mean squared error corresponding to the output ofthe equalizer. As shown in FIG. 26, when equalization is performed inthe frequency domain, the distortion of data caused by a multiple pathmay be compensated. However, a white noise element that is added to theoutput of the multiple path channel is amplified by the frequency domainequalizer, thereby changing to a colored noise. By removing the colorednoise that has been amplified as described above, the equalizationperformance of the frequency domain channel equalizer may be enhanced.

FIG. 27 and FIG. 28 illustrate exemplary structures of a channelequalizer that is used for resolving noise amplification problems causedby the frequency domain equalizer as described above. FIG. 27illustrates a block diagram showing the channel equalization devicehaving a time domain equalizer 1100 added to the frequency domainequalizer shown in FIG. 26. Herein, the time domain equalizer 1100includes an adder 1101, a decision device 1102, and a feedback filter1103. More specifically, the data that are equalized by the frequencydomain equalizer 1000 and converted to time domain data, as shown inFIG. 26, are inputted to the adder 1101 of the time domain equalizer1100. The adder 1101 then adds the output of the frequency domainequalizer 1000 and the output of the feedback filter 1103. Thereafter,the added result is outputted for data recovery and simultaneouslyoutputted to the decision device 1102.

The decision device 1102 compares the output signal outputted from adder1101 with a predetermined reference signal level. Accordingly, thereference signal level that is the nearest to the output signal of theadder 1101 is decided and outputted as the decision value. Subsequently,the feedback filter 1103 receives the decision value of the decisiondevice 1102, so as to perform a filtering process in the time domain.Then, the processed decision value is outputted to the adder 1101. Atthis point, if the decision is accurately made by the decision device1102, since the decision value having the noise removed (or cancelled)from the output element of the frequency domain equalizer 1000 isre-inputted as the input of the feedback filter 1103, amplification ofthe noise does not occur. Therefore, the equalization performance of thechannel equalizer of FIG. 27 is more excellent than the channelequalizer of FIG. 26.

FIG. 28 illustrates a block diagram showing the channel equalizationdevice having a noise canceller 1200 added to the frequency domainequalizer 1000 shown in FIG. 26. Herein, the noise canceller 1200includes a subtracter 1201, a decision device 1202, and a noisepredictor 1203. More specifically, the data that have been equalized bythe frequency domain channel equalizer 1000 shown in FIG. 26 andconverted to time domain data are inputted to the subtracter 1201 of thenoise canceller 1200. The subtracter 1201 removes the noise, which ispredicted by the noise predictor 1203, from the signal equalized in thefrequency domain. Then, the signal having the amplified noise removedtherefrom is outputted for data recovery and simultaneously outputted tothe decision device 1202.

The decision device 1202 outputs a decision value closest to the outputof the subtracter 1201 to the noise predictor 1203. The noise predictor1203 removes the output of the decision device 1202 from the signal thathas been converted to the time domain by the frequency domain equalizer1000, so as to calculate the noise element. Thereafter, the calculatednoise element is used as an input of a filter (not shown) within thenoise predictor 1203. The noise predictor 1203 uses the filter topredict the colored noise element included in the output symbol of thecurrent frequency domain equalizer 1000. Then, the predicted colorednoise element is outputted to the subtracter 1201. Subsequently, thesubtracter 1201 removes the noise element predicted by the noisepredictor 1203 from the output of the frequency domain equalizer 1000,thereby outputting the final data. As described above, by periodicallyinserting and transmitting general data in known data having identicalpatterns, and by having the receiving system perform channelequalization in the form of circular convolution, the structure of thechannel equalizer according to the present invention becomes moresimplified, and the channel equalizing performance may be enhanced.

Meanwhile, as described above, when the input data are converted tofrequency domain data by the first FFT unit 1001, the FFT block sectionshould be configured so that part of the known data of the known datasection is located both in front of and behind the general data section,as shown in FIG. 23. Thus, the influence of the channel may occur in theform of circular convolution, thereby allowing channel equalization tobe performed correctly. However, when equalization is performed by thechannel equalizer shown in FIG. 26 to FIG. 28, and when the CIR of pointT1 and the CIR of point T2 (both shown in FIG. 23) are used for theequalization process, the point of the general data to which the actualequalization is applied and the point of the CIR used in theequalization process do not coincide with one another. This leads to adeficiency in the performance of the equalizer. More specifically, thisis because the CIR is the value calculated (or obtained) from the knowndata section, and because the channel equalization process of thegeneral data section is performed by using the CIR calculated (orobtained) from the known data section.

In order to resolve such problems, another example of an indirectequalization type channel equalizer is presented in FIG. 29. Herein, anaverage value of the CIRs is used for channel equalizing the generaldata. For example, when performing the equalization process, an averagevalue of the CIR of point T1 and the CIR of point T2 is used to performa channel equalization process on the general data section between pointT1 and point T2, thereby enhancing the equalizing performance. In thiscase, each known data section may use the CIR obtained from eachcorresponding known data section so as to perform channel equalization.As another example, the channel equalization may be performed in theknown data section including point T1, the known data section includingpoint T2, and a general data section between point T1 and point T2 byusing the average CIR value between point T1 and point T2. Accordingly,the channel equalizer shown in FIG. 29 further includes a first FFT unit1301, a distortion compensator 1302, a CIR estimator 1303, an averagecalculator 1304, a second FFT unit 1305, a coefficient calculator 1306,and an inverse IFFT unit 1307. In the structure shown in FIG. 29, theoperations of the first FFT unit 1301, the distortion compensator 1302,the CIR estimator 1303, and the IFFT unit 1307 may be identical to thoseshown in FIG. 26.

More specifically, the CIR estimator 1303 estimates the CIR by using thedata received during the known data section and the known data sequence.Then, the estimated CIR is outputted to the average calculator 1304.Herein, the average calculator 1304 calculates and obtains an averagevalue of the consecutive CIRs that are being inputted and, then, outputsthe calculated average value to the second FFT unit 1305. For example,the average value between the CIR value estimated at point T1 and theCIR value estimated at point T2 is calculated, so that the calculatedaverage value can be used in the channel equalization process of thegeneral data located between point T1 and point T2. Accordingly, thecalculated average value is outputted to the second FFT unit 1305. Thesecond FFT unit 1305 converts the estimated CIR to the frequency domainCIR, which is then outputted to the coefficient calculator 1306. Thecoefficient calculator 1306 uses the average CIR of the frequency domainin order to calculate the equalization coefficient, thereby outputtingthe calculated coefficient to the distortion compensator 1302. Herein,for example, the coefficient calculator 1306 may obtain and to outputthe calculated the equalization coefficient domain that can provideminimum mean square error (MMSE) from the estimated average CIR of thefrequency domain. The operations of the components that follow areidentical to those shown in FIG. 26, and so the description of the samewill, therefore, be omitted for simplicity.

FIG. 30 illustrates a block diagram of a channel equalizer according toanother embodiment of the present invention. More specifically, FIG. 30illustrates a block diagram of a frequency domain channel equalizerusing an indirect equalization method according to another embodiment ofthe present invention. Herein, an overlap & save method is used toperform linear convolutional operation in the frequency domain.Referring to FIG. 30, the channel equalizer includes an overlap unit1401, a first fast fourier transform (FFT) unit 1402, a distortioncompensator 1403, a CIR estimator 1404, a second FFT unit 1405, acoefficient calculator 1406, an inverse fast fourier transform (IFFT)unit 1407, and a save unit 1408. Herein, any device performing complexnumber multiplication may be used as the distortion compensator 1403.

In the channel equalizer having the above-described structure, as shownin FIG. 30, the received data are overlapped by the overlap unit 1401and then inputted to the first FFT unit 1402. The first FFT unit 1402converts (or transforms) the overlapped data of the time domain tooverlapped data of the frequency domain by using fast fourier transform(FFT). Then, the converted data are outputted to the distortioncompensator 1403. The distortion compensator 1403 performs complexmultiplication on the equalization coefficient calculated from thecoefficient calculator 1406 and the overlapped data of the frequencydomain, thereby compensating the channel distortion of the overlappeddata being outputted from the first FFT unit 1402. Thereafter, thedistortion-compensated data are outputted to the IFFT unit 1407. TheIFFT unit 407 performs inverse fast fourier transform (IFFT) on thedistortion-compensated overlapped data, so as to convert thecorresponding data back to data (i.e., overlapped data) of the timedomain. Thereafter, the converted data are outputted to the save unit1408. The save unit 1408 extracts only the valid data from theoverlapped data of the time domain, thereby outputting the extractedvalid data.

Meanwhile, the CIR estimator 1404 estimates the CIR by using the datareceived during the known data section and the known data of the knowndata section, the known data being already known by the receiving systemin accordance with an agreement between the receiving system and thetransmitting system. Then, the estimated CIR is outputted to the secondFFT unit 1405. For this, the CIR estimator 1404 receives known sequenceposition indicators from the known data detector 904. Furthermore, sincethe known data sequence of the known data section (or block) correspondsto data already known according to an agreement between the receivingsystem and the transmitting system, the known data sequence may bestored in advance (or pre-stored) in the CIR estimator 1404.Alternatively, the known data sequences generated from the known datadetector 904 may also be received.

The second FFT unit 1405 performs FFT on the estimated CIR of the timedomain in order to convert the time domain CIR to a frequency domainCIR. Then, the second FFT unit 1405 outputs the converted CIR to thecoefficient calculator 1406.

The coefficient calculator 1406 uses the estimated CIR of the frequencydomain, so as to calculate the equalization coefficient and to outputthe calculated equalization coefficient to the distortion compensator1403. The coefficient calculator 1406 is an exemplary part of thechannel equalizer according to the embodiment of the present invention.At this point, the coefficient calculator 1406 calculates a channelequalization coefficient of the frequency domain that can provideminimum mean square error (MMSE) from the estimated CIR of the frequencydomain. The calculated channel equalization coefficient is outputted tothe distortion compensator 1403. More specifically, the channelequalizer of FIG. 26 performs circular convolution calculation in thefrequency domain in order to perform the channel equalization process.On the other hand, the frequency domain channel equalizer of FIG. 30performs linear convolution calculation in the frequency domain in orderto perform the channel equalization process. Other than this, theremaining components and the corresponding operations are identical toone another and so, a detailed description of the same will, therefore,be omitted for simplicity.

Since the channel equalizer of FIG. 30 does not use the characteristicsof the guard interval, unlike in FIG. 26, the channel equalizer shown inFIG. 30 is advantageous in that there are no limitations whendetermining the FFT block section. More specifically, the known data donot necessarily have to be located in front of and behind the generaldata which are inputted to the first FFT unit FTT1. The channelequalizers are used to estimate the CIRs by using the known data thatare periodically transmitted. Since the CIRs are used to calculate eachequalization coefficient, the speed at which the equalizationcoefficient is updated largely depends the cycle period according towhich the known data are being transmitted. Therefore, in a dynamicchannel (i.e., in a channel wherein the characteristics change inaccordance with the time), when the channel changing speed is fasterthan the transmission cycle of the known data, the equalizationperformance may be deteriorated. At this point, when the known data arefrequently transmitted in order to compensate the channel that undergoesfrequent and fast changes, the transmission efficiency of the generaldata, in which the actual valid contents are transmitted, may bedeteriorated. Accordingly, there are limitations in reducing thetransmission cycle of the known data.

In order to resolve such problems, another example of a channelequalizer is presented in FIG. 31. FIG. 31 illustrates a block diagramof a channel equalizer of the frequency domain according to anotherembodiment of the present invention. Instead of frequently transmittingthe known data, the channel equalizer according to another embodiment ofthe present invention interpolates CIRs that are estimated in the knowndata section and uses the interpolated CIRs for the channel equalizationof the general data. Herein, the channel equalizer of FIG. 31 furtherincludes an interpolator 1409 between the CIR estimator 1404 and thesecond FFT unit 1405 of FIG. 30.

More specifically, the CIR estimator 1404 estimates the CIR by using thedata received during the known data section and the known data of theknown data section already known by the receiving system according to anagreement between the receiving system and the transmitting system.Then, the estimated CIR is outputted to the interpolator 1409. Herein,the interpolator 1409 uses the inputted CIR to estimate CIR of thesection that does not include the known data by using a pre-determinedinterpolation method. Then, the interpolator 1409 outputs the estimatedCIR to the second FFT unit 1405. The second FFT unit 1405 converts theinputted CIR to the frequency domain CIR, which is then outputted to thecoefficient calculator 1406. The coefficient calculator 1406 uses theinterpolated CIR of the frequency domain in order to calculate theequalization coefficient that can provide minimum mean square error(MMSE) and to output the calculated equalization coefficient to thedistortion compensator 1403. The operations of the components thatfollow are identical to those shown in FIG. 30, and so the descriptionof the same will, therefore, be omitted for simplicity.

Herein, the interpolation method of the interpolator 1409 corresponds toa method of estimating data of an unknown point by using the data knownfrom a particular function. The simplest example of interpolation is thelinear interpolation, which is illustrated in FIG. 32. Morespecifically, in a random function F(x), when given the values F(Q) andF(S) each from points x=Q and x=S, respectively, the estimated value{circumflex over (F)}(P) of the F(x) function at point x=P may beestimated by using Equation 4 below.

$\begin{matrix}{{{\hat{F}(P)} = {{\frac{{F(S)} - {F(Q)}}{S - Q}\left( {P - Q} \right)} + {F(Q)}}}{{\hat{F}(P)} = {{\frac{S - P}{S - Q}{F(Q)}} + {\frac{P - Q}{S - Q}{F(S)}}}}} & {{Equation}\mspace{14mu} 4}\end{matrix}$

If the known data sequence is periodically inserted and transmitted, asshown in FIG. 23, the distance between the general data section and theknown data section positioned in front of or behind the general datasection, according to the embodiment of the present invention, isdetermined as one cycle period. Herein, at least one of the datasections included in the one cycle period is divided into a plurality ofsections. The CIR is estimated in each divided section by using theabove-described interpolation method. However, when one data cycle isdivided into a plurality of sections, not all FFT block sections can beconfigured to have the known data located (or positioned) before orbehind the FFT block section, as shown in FIG. 23. Therefore, theabove-described interpolation method cannot be used in the channelequalizer using the cyclic prefix. Nevertheless, when using the channelequalizer using the overlap & save method, as shown in FIG. 31.

FIG. 33 illustrates an example of overlap & save processes according toan embodiment of the present invention, when linearly interpolating theCIR. Referring to FIG. 33, data are overlapped so that the current FFTblock and the previous FFT block are overlapped at an overlapping ratioof 50%, and the data cycle period including the general data section andthe known data section positioned in front of the general data sectionis divided into 4 sections. In this example, the size and number of eachsection are decided so that a known data sequence is included in atleast one section of the data cycle. This is to enable FFT block data ofthe section including the known data sequence directly to use the CIRestimated from the corresponding section for the channel equalizationprocess. Furthermore, the CIR obtained (or calculated) from the datasection including the known data sequence is used for interpolating theCIR of the section that does not include any known data sequence.

Herein, the data inputted to the first FFT unit 1402 are inserted inorder from FFT block 1 to FFT block 5, as shown in FIG. 33. The inputteddata are then inputted to the distortion compensator 1403 so as to bechannel equalized. Subsequently, the equalized data are converted totime domain data by the IFFT unit 1407. Then, only the portioncorresponding to valid data is extracted from the converted by the saveunit 1408. Thereafter, the final data are outputted.

Referring to FIG. 33, sections 1 to 5 represent the valid data sectioncorresponding to each FFT block. At this point, since a known datasequence is included in the FFT block 1, the CIR T2 may be estimated.Similarly, since a known data sequence is included in the FFT block 5,the CIR T3 may also be estimated. In other words, the data of the FFTblock 1 use the estimated CIR T2 to perform channel equalization, andthe data of the FFT block 5 uses the estimated CIR T3 to perform channelequalization. However, since the known data sequence is not included inFFT block 2 to FFT block 4, the interpolator 1409 interpolates CIR T2and CIR T3, thereby estimating the CIRs that may each be used in the FFTblock 2 to the FFT block 4. At this point, if linear interpolation isused, the CIRs corresponding to FFT block 2 to FFT block 4 may beestimated by using Equation 5 shown below.

FFT Block 2:0.75T2+0.25T3

FFT Block 3:0.5T2+0.5T3

FFT Block 4:0.25T2+0.75T3  Equation 5

Referring to FIG. 34, data are overlapped so that the current FFT blockand the previous FFT block are overlapped at an overlapping ratio of75%, and the data cycle period including the general data section andthe known data section positioned in front of the general data sectionis divided into 4 sections. Similarly, in this example, the size andnumber of each section are decided so that a known data sequence isincluded in at least one section of the data cycle. Furthermore, theinterpolated and estimated CIRs are identical as those shown in FIG. 33.Yet, the only difference is that the FFT block size is twice the size ofthe FFT blocks of FIG. 33. As described above, the examples shown inFIG. 33 and FIG. 34 are merely exemplary, and the degree and algorithmof the interpolation method and the size of the FFT block may be setdiversely whenever required or necessary.

Herein, the data group that is inputted for the equalization process isdivided into regions A to C, as shown in FIG. 6A. More specifically, inthe example of the present invention, each region A, B, and C arefurther divided into regions A1 to A5, regions B1 and B2, and regions C1to C3, respectively. Most particularly, an example of estimating the CIRin accordance with each region A (A1 to A5), B (B1 and B2), and C (C1 toC3) within the data group, which is hierarchically divided andtransmitted from the transmitting system, and applying each CIRdifferently will also be described herein. Furthermore, by using theknown data, the place and contents of which is known in accordance withan agreement between the transmitting system and the receiving system,and the field synchronization data, so as to estimate the CIR, thepresent invention may be able to perform channel equalization with morestability.

Referring to FIG. 6A, the CIR that is estimated from the fieldsynchronization data in the data structure is referred to as CIR_FS.Alternatively, the CIRs that are estimated from each of the 5 known datasequences existing in region A are sequentially referred to as CIR_N0,CIR_N1, CIR_N2, CIR_N3, and CIR_N4. As described above, the presentinvention uses the CIR estimated from the field synchronization data andthe known data sequences in order to perform channel equalization ondata within the data group. At this point, each of the estimated CIRsmay be directly used in accordance with the characteristics of eachregion within the data group. Alternatively, a plurality of theestimated CIRs may also be either interpolated or extrapolated so as tocreate a new CIR, which is then used for the channel equalizationprocess.

Herein, as shown in FIG. 32 to FIG. 34, when a value F(Q) of a functionF(x) at a particular point Q and a value F(S) of the function F(x) atanother particular point S are known, interpolation refers to estimatinga function value of a point within the section between points Q and S.Linear interpolation corresponds to the simplest form among a wide rangeof interpolation operations. The linear interpolation described hereinis merely exemplary among a wide range of possible interpolationmethods. And, therefore, the present invention is not limited only tothe examples set forth herein. Alternatively, when a value F(Q) of afunction F(x) at a particular point Q and a value F(S) of the functionF(x) at another particular point S are known, extrapolation refers toestimating a function value of a point outside of the section betweenpoints Q and S. Linear extrapolation is the simplest form among a widerange of extrapolation operations. Similarly, the linear extrapolationdescribed herein is merely exemplary among a wide range of possibleextrapolation methods.

FIG. 35 illustrates an example of an extrapolation operation.Particularly, FIG. 35 illustrates the simplest form of linearextrapolation among a wide range of extrapolation operations. Morespecifically, since the values F(Q) and F(S) each from points Q and S,respectively, are known, a straight line passing between the Q-S sectionmay be obtained so as to calculate an approximate value F(P) of the F(x)function at point P. Herein, the equation of the straight line passingthe coordinates (Q,F(Q)) and (S,F(S)) is shown in Equation 6 below:

$\begin{matrix}{{\hat{F}(x)} = {{\frac{{F(S)} - {F(Q)}}{S - Q}\left( {x - Q} \right)} + {F(Q)}}} & {{Equation}\mspace{14mu} 6}\end{matrix}$

Therefore, in order to calculate the function value at point P usingEquation 6, the equation shown in Equation 7 below may be used.

$\begin{matrix}{{\hat{F}(P)} = {{\frac{{F(S)} - {F(Q)}}{S - Q}\left( {P - Q} \right)} + {F(Q)}}} & {{Equation}\mspace{14mu} 7}\end{matrix}$

The above-described linear extrapolation method is the simplest form oflinear extrapolation among a wide range of extrapolation operations.However, the method described above is merely exemplary, and, therefore,the present invention is not limited only to the example presentedabove. Accordingly, a variety of other extrapolation may be used in thepresent invention.

FIG. 36 illustrates a conceptual diagram of the extrapolation operationaccording to the present invention. Most particularly, in the example ofthe extrapolation operation shown in FIG. 36, data of a current FFTblock and data of a previous FFT block are overlapped at an overlappingratio of 50%, then regions B and C are divided into a plurality ofsections so that the CIR of each section is estimated by performing (oroperating) extrapolation on the CIRs of region A. More specifically, theCIR of each section of regions B and C is estimated by performingextrapolation on at least one of the CIRs of region A. Then, by usingeach of the estimated CIRs, channel equalization is performed on thedata corresponding to each section. When extrapolation is performed onCIR_N3 and CIR_N4 of region A, in order to estimate the CIRs of section1 to section 4 of regions B and C, and when the length of one section isassumed to be equal to 4 segments, the CIRs of section 1 to section 4having linear extrapolation performed thereon are as follows:

-   -   Section 1: 1.25*CIR_N4−0.25*CIR_N3    -   Section 2: 1.5*CIR_N4−0.5*CIR_N3    -   Section 3: 1.75*CIR_N4−0.75*CIR_N3    -   Section 4: 2*CIR_N4−CIR_N3

More specifically, in case of region C1, any one of the CIR_N4 estimatedfrom a previous data group, the CIR_FS estimated from the current datagroup that is to be processed with channel equalization, and a new CIRgenerated by extrapolating the CIR_FS of the current data group and theCIR_N0 may be used to perform channel equalization. Alternatively, incase of region B1, a variety of methods may be applied as described inthe case for region C1. For example, a new CIR created by linearlyextrapolating the CIR_FS estimated from the current data group and theCIR_N0 may be used to perform channel equalization. Also, the CIR_FSestimated from the current data group may also be used to performchannel equalization. Finally, in case of region A1, a new CIR may becreated by interpolating the CIR_FS estimated from the current datagroup and CIR_N0, which is then used to perform channel equalization.Furthermore, any one of the CIR_FS estimated from the current data groupand CIR_N0 may be used to perform channel equalization.

In case of regions A2 to A5, CIR_N(i−1) estimated from the current datagroup and CIR_N(i) may be interpolated to create a new CIR and use thenewly created CIR to perform channel equalization. Also, any one of theCIR_N(i−1) estimated from the current data group and the CIR_N(i) may beused to perform channel equalization. Alternatively, in case of regionsB2, C2, and C3, CIR_N3 and CIR_N4 both estimated from the current datagroup may be extrapolated to create a new CIR, which is then used toperform the channel equalization process. Furthermore, the CIR_N4estimated from the current data group may be used to perform the channelequalization process. Accordingly, an optimum performance may beobtained when performing channel equalization on the data inserted inthe data group.

FIG. 37 illustrates a block diagram of a channel equalizer according toanother embodiment of the present invention. Referring to FIG. 37, thechannel equalizer includes an overlap unit 1501, a first fast fouriertransform (FFT) unit 1502, a distortion compensator 1503, a CIRestimator 1504, a first pre-CIR cleaner 1505, a CIR calculator 1506, asecond pre-CIR cleaner 1507, a zero-padding unit 1508, a second FFT unit1509, a coefficient calculator 1510, an inverse fast fourier transform(IFFT) unit 1511, and a save unit 1512. Herein, any device that performscomplex number multiplication may be used as the distortion compensator1503.

In the channel equalizer having the above-described structure, as shownin FIG. 37, the received data are overlapped by the overlap unit 1501and are outputted to the first FFT unit 1502. The first FFT unit 1502performs a FFT process to convert the overlapped data of the time domainto overlapped data of the frequency domain. Then, the convertedoverlapped data are outputted to distortion compensator 1503. Thedistortion compensator 1503 performs complex multiplication onoverlapped data of the frequency domain and the equalization coefficientcalculated by the coefficient calculator 1510, thereby compensating thechannel distortion of the overlapped data being outputted from the firstFFT unit 1502. Thereafter, the distortion-compensated data are outputtedto the IFFT unit 1511. The IFFT unit 1511 performs inverse fast fouriertransform (IFFT) on the distortion-compensated overlapped data, so as toconvert the corresponding data back to data (i.e., overlapped data) ofthe time domain. Then, the converted data are outputted to the save unit1512. The save unit 1512 extracts and outputs only the valid data fromthe overlapped data of the time domain.

Meanwhile, the CIR estimator 1504 uses the data being received duringthe known data section and the known data of the known data sectionalready known by the receiving system in accordance with an agreementbetween the receiving system and the transmitting system in order toestimate to the channel impulse response (CIR). For this, the CIRestimator 1504 receives known sequence position indicators from theknown data detector 904. Also, since the known data of the known datasection correspond to data already known in accordance with an agreementbetween the receiving system and the transmitting system, the known datamay be pre-stored in the CIR estimator 1504, or the known data generatedfrom the known data detector 904 may also be received. The CIR estimator1504 may estimate the CIR by using a least square (LS) method.

The CIR estimated by the CIR estimator 1504 either passes through thefirst pre-CIR cleaner 1505 or bypasses the first pre-CIR cleaner 1505,thereby being inputted to the CIR calculator 1506. The CIR calculator1506 then performs interpolation or extrapolation on the estimated CIR,which is then outputted to the second pre-CIR cleaner 1507. Herein,depending upon whether the CIR calculator 1506 performs interpolation orextrapolation on the estimated CIR, the first pre-CIR cleaner 1505 orthe second pre-CIR cleaner 1507 may be operated. For example, if the CIRcalculator 1506 performs interpolation on the estimated CIR, the secondpre-CIR cleaner 1507 is operated. Alternatively, if the CIR calculator1506 extrapolates the estimated CIR, the first pre-CIR cleaner 1505 isoperated.

More specifically, the Cir estimated from the known data not onlyincludes the channel element that is to be obtained, but also includesjitter elements caused by noise. Such jitter element results in adeterioration of the performance of the channel equalizer. Accordingly,it is preferable to remove the jitter element before using the CIR inthe coefficient calculator 1510. Therefore, in the examples givenherein, the first and second pre-CIR cleaners 1505 and 1507 respectivelyremove portions of the CIR element having a power level equal to andlower than a pre-determined threshold value from the CIR element (i.e.,the first and second pre-CIR cleaners 1505 and 1507 respectively processthe CIR element to ‘0’). Herein, the above-described process is referredto as a CIR cleaning process.

In other words, in the CIR calculator 1506, the CIR interpolationprocess is performed by multiplying and adding coefficients to each ofthe two CIR estimated by the CIR estimator 1504. At this point, aportion of the noise element of each CIR is added to one another,thereby offsetting (or canceling) one another. Therefore, when the CIRcalculator 1506 performs CIR interpolation, the original CIR having thenoise element remaining therein. More specifically, when the CIRcalculator 1506 performs CIR interpolation, the CIR estimated by the CIRestimator 1504 bypasses the first pre-CIR cleaner 1505 and then inputtedto the CIR calculator 1506. Then, the CIR interpolated by the CIRcalculator 1506 is processed with CIR cleaning by the second pre-CIRcleaner 1507.

Conversely, in the CIR calculator 1506, the CIR extrapolation process isperformed by using a difference value between two CIRS estimated by theCIR estimator 1504 so as to estimate a CIR located outside of the twoestimated CIRs. In this case, the noise element is amplified. Therefore,when the CIR calculator 1506 performs CIR extrapolation, the CIRprocessed with CIR cleaning by the first pre-CIR cleaner 1505 is used.More specifically, when the CIR calculator 1506 performs CIRextrapolation, the extrapolated CIR bypasses the second pre-CIR cleaner1507 and is inputted to the zero-padding unit 1508.

Meanwhile, when the second FFT unit 1509 converts the CIR that eitherbypasses the second pre-CIR cleaner 1507 or is processed with CIRcleaning to a frequency domain CIR, the length of the CIR being inputtedmay not be equal to the FFT size. In other words, the length of the CIRmay be shorter than the FFT size. In this case, the zero-padding unit1508 adds to the CIR the same amount of zeros (0) corresponding todifference between the FFT size and the length of the CIR beinginputted. Thereafter, the processed data are outputted to the second FFTunit 1509. Herein, the CIR processed with zero-padding may correspond toany one of the interpolated CIR, the extrapolated CIR, and the CIRestimated from the known data section.

The second FFT unit 1509 performs FFT on the CIR of the time domain,which is being inputted, so as to convert the time domain CIR to afrequency domain CIR, which is then outputted to the coefficientcalculator 1510. The coefficient calculator 1505 then uses the convertedfrequency domain CIR to calculate the equalization coefficient, therebyoutputting the calculated coefficient to the distortion compensator1503. At this point, the coefficient calculator 1503 calculates achannel equalization coefficient of the frequency domain that canprovide minimum mean square error (MMSE) from the frequency domain CIR.Then, the calculated coefficient may be outputted to the distortioncompensator 1503.

FIG. 38 illustrates a block diagram of a channel equalizer according toanother embodiment of the present invention. In other words, FIG. 38illustrates a block diagram showing another example of a channelequalizer by using different CIR estimation and application methods inaccordance with regions A, B, and C, when the data group is divided intothe structure shown in FIG. 6A. More specifically, as shown in FIG. 6A,known data that are sufficiently are being periodically transmitted inregion A. Therefore, an indirect equalizing method using the CIR may beused herein. However, in regions B and C, the known data are neitherable to be transmitted at a sufficiently long length nor able to beperiodically and equally transmitted. Therefore, it is inadequate toestimate the CIR by using the known data. Accordingly, in regions B andC, a direct equalizing method in which an error is obtained from theoutput of the equalizer, so as to update the coefficient. The examplespresented in the embodiments of the present invention shown in FIG. 38include a method of performing indirect channel equalization by using acyclic prefix on the data of region A, and a method of performing directchannel equalization by using an overlap & save method on the data ofregions B and C.

Accordingly, referring to FIG. 38, the frequency domain channelequalizer includes a frequency domain converter 1610, a distortioncompensator 1620, a time domain converter 1630, a first coefficientcalculating unit 1640, a second coefficient calculating unit 1650, and acoefficient selector 1660. Herein, the frequency domain converter 1610includes an overlap unit 1611, a select unit 1612, and a first FFT unit1613. The time domain converter 1630 includes an IFFT unit 1631, a saveunit 1632, and a select unit 1633. The first coefficient calculatingunit 1640 includes a CIR estimator 1641, an average calculator 1642, andsecond FFT unit 1643, and a coefficient calculator 1644. The secondcoefficient calculating unit 1650 includes a decision unit 1651, aselect unit 1652, a subtractor 1653, a zero-padding unit 1654, a thirdFFT unit 1655, a coefficient updater 1656, and a delay unit 1657. Also,a multiplexer (MUX), which selects data that are currently beinginputted as the input data depending upon whether the data correspond toregion A or to regions B/C, may be used as the select unit 1612 of thefrequency domain converter 1610, the select unit 1633 of the time domainconverter 1630, and the coefficient selector 1660.

In the channel equalizer having the above-described structure, as shownin FIG. 38, if the data being inputted correspond to the data of regionA, the select unit 1612 of the frequency domain converter 1610 selectsthe input data and not the output data of the overlap unit 1611. In thesame case, the select unit 1633 of the time domain converter 1630selects the output data of the IFFT unit 1631 and not the output data ofthe save unit 1632. The coefficient selector 1660 selects theequalization coefficient being outputted from the first coefficientcalculating unit 1640. Conversely, if the data being inputted correspondto the data of region A, the select unit 1612 of the frequency domainconverter 1610 selects the output data of the overlap unit 1611 and notthe input data. In the same case, the select unit 1633 of the timedomain converter 1630 selects the output data of the save unit 1632 andnot the output data of the IFFT unit 1631. The coefficient selector 1660selects the equalization coefficient being outputted from the secondcoefficient calculating unit 1650.

More specifically, the received data are inputted to the overlap unit1611 and select unit 1612 of the frequency domain converter 1610, and tothe first coefficient calculating unit 1640. If the inputted datacorrespond to the data of region A, the select unit 1612 selects thereceived data, which are then outputted to the first FFT unit 1613. Onthe other hand, if the inputted data correspond to the data of regions Band C, the select unit 1612 selects the data that are overlapped by theoverlap unit 1613 and are, then, outputted to the first FFT unit 1613.The first FFT unit 1613 performs FFT on the time domain data that areoutputted from the select unit 1612, thereby converting the time domaindata to frequency domain data. Then, the converted data are outputted tothe distortion compensator 1620 and the delay unit 1657 of the secondcoefficient calculating unit 1650.

The distortion compensator 1620 performs complex multiplication onfrequency domain data outputted from the first FFT unit 1613 and theequalization coefficient outputted from the coefficient selector 1660,thereby compensating the channel distortion detected in the data thatare being outputted from the first FFT unit 1613. Thereafter, thedistortion-compensated data are outputted to the IFFT unit 1631 of thetime domain converter 1630. The IFFT unit 1631 of the time domainconverter 1630 performs IFFT on the channel-distortion-compensated data,thereby converting the compensated data to time domain data. Theconverted data are then outputted to the save unit 1632 and the selectunit 1633. If the inputted data correspond to the data of region A, theselect unit 1633 selects the output data of the IFFT unit 1631. On theother hand, if the inputted data correspond to regions B and C, theselect unit 1633 selects the valid data extracted from the save unit1632. Thereafter, the selected data are outputted to be decoded and,simultaneously, outputted to the second coefficient calculating unit1650.

The CIR estimator 1641 of the first coefficient calculating unit 1640uses the data being received during the known data section and the knowndata of the known data section, the known data being already known bythe receiving system in accordance with an agreement between thereceiving system and the transmitting system, in order to estimate theCIR. Subsequently, the estimated CIR is outputted to the averagecalculator 1642. The average calculator 1642 calculates an average valueof the CIRs that are being inputted consecutively. Then, the calculatedaverage value is outputted to the second FFT unit 1643. For example,referring to FIG. 23, the average value of the CIR value estimated atpoint T1 and the CIR value estimated at point T2 is used for the channelequalization process of the general data existing between point T1 andpoint T2. Accordingly, the calculated average value is outputted to thesecond FFT unit 1643.

The second FFT unit 1643 performs FFT on the CIR of the time domain thatis being inputted, so as to convert the inputted CIR to a frequencydomain CIR. Thereafter, the converted frequency domain CIR is outputtedto the coefficient calculator 1644. The coefficient calculator 1644calculates a frequency domain equalization coefficient that satisfiesthe condition of using the CIR of the frequency domain so as to minimizethe mean square error. The calculated equalizer coefficient of thefrequency domain is then outputted to the coefficient calculator 1660.

The decision unit 1651 of the second coefficient calculating unit 1650selects one of a plurality of decision values (e.g., 8 decision values)that is most approximate to the equalized data and outputs the selecteddecision value to the select unit 1652. Herein, a multiplexer may beused as the select unit 1652. In a general data section, the select unit1652 selects the decision value of the decision unit 1651.Alternatively, in a known data section, the select unit 1652 selects theknown data and outputs the selected known data to the subtractor 1653.The subtractor 1653 subtracts the output of the select unit 1633included in the time domain converter 1630 from the output of the selectunit 652 so as to calculate (or obtain) an error value. Thereafter, thecalculated error value is outputted to the zero-padding unit 1654.

The zero-padding unit 1654 adds (or inserts) the same amount of zeros(0) corresponding to the overlapped amount of the received data in theinputted error. Then, the error extended with zeros (0) is outputted tothe third FFT unit 1655. The third FFT unit 1655 converts the error ofthe time domain having zeros (0) added (or inserted) therein, to theerror of the frequency domain. Thereafter, the converted error isoutputted to the coefficient update unit 1656. The coefficient updateunit 1656 uses the received data of the frequency domain that have beendelayed by the delay unit 1657 and the error of the frequency domain soas to update the previous equalization coefficient. Thereafter, theupdated equalization coefficient is outputted to the coefficientselector 1660.

At this point, the updated equalization coefficient is stored so as thatit can be used as a previous equalization coefficient in a laterprocess. If the input data correspond to the data of region A, thecoefficient selector 1660 selects the equalization coefficientcalculated from the first coefficient calculating unit 1640. On theother hand, if the input data correspond to the data of regions B and C,the coefficient selector 1660 selects the equalization coefficientupdated by the second coefficient calculating unit 1650. Thereafter, theselected equalization coefficient is outputted to the distortioncompensator 1620.

FIG. 39 illustrates a block diagram of a channel equalizer according toanother embodiment of the present invention. In other words, FIG. 39illustrates a block diagram showing another example of a channelequalizer by using different CIR estimation and application methods inaccordance with regions A, B, and C, when the data group is divided intothe structure shown in FIG. 6A. In this example, a method of performingindirect channel equalization by using an overlap & save method on thedata of region A, and a method of performing direct channel equalizationby using an overlap & save method on the data of regions B and C areillustrated.

Accordingly, referring to FIG. 39, the frequency domain channelequalizer includes a frequency domain converter 1710, a distortioncompensator 1720, a time domain converter 1730, a first coefficientcalculating unit 1740, a second coefficient calculating unit 1750, and acoefficient selector 1760. Herein, the frequency domain converter 1710includes an overlap unit 1711 and a first FFT unit 1712. The time domainconverter 1730 includes an IFFT unit 1731 and a save unit 1732. Thefirst coefficient calculating unit 1740 includes a CIR estimator 1741,an interpolator 1742, a second FFT unit 1743, and a coefficientcalculator 1744. The second coefficient calculating unit 1750 includes adecision unit 1751, a select unit 1752, a subtractor 1753, azero-padding unit 1754, a third FFT unit 1755, a coefficient updater1756, and a delay unit 1757. Also, a multiplexer (MUX), which selectsdata that are currently being inputted as the input data depending uponwhether the data correspond to region A or to regions B and C, may beused as the coefficient selector 1760. More specifically, if the inputdata correspond to the data of region A, the coefficient selector 1760selects the equalization coefficient calculated from the firstcoefficient calculating unit 1740. On the other hand, if the input datacorrespond to the data of regions B and C, the coefficient selector 1760selects the equalization coefficient updated by the second coefficientcalculating unit 1750.

In the channel equalizer having the above-described structure, as shownin FIG. 39, the received data are inputted to the overlap unit 1711 ofthe frequency domain converter 1710 and to the first coefficientcalculating unit 1740. The overlap unit 1711 overlaps the input data toa pre-determined overlapping ratio and outputs the overlapped data tothe first FFT unit 1712. The first FFT unit 1712 performs FFT on theoverlapped time domain data, thereby converting the overlapped timedomain data to overlapped frequency domain data. Then, the converteddata are outputted to the distortion compensator 1720 and the delay unit1757 of the second coefficient calculating unit 1750.

The distortion compensator 1720 performs complex multiplication on theoverlapped frequency domain data outputted from the first FFT unit 1712and the equalization coefficient outputted from the coefficient selector1760, thereby compensating the channel distortion detected in theoverlapped data that are being outputted from the first FFT unit 1712.Thereafter, the distortion-compensated data are outputted to the IFFTunit 1731 of the time domain converter 1730. The IFFT unit 1731 of thetime domain converter 1730 performs IFFT on the distortion-compensateddata, thereby converting the compensated data to overlapped time domaindata. The converted overlapped data are then outputted to the save unit1732. The save unit 1732 extracts only the valid data from theoverlapped time domain data, which are then outputted for data decodingand, at the same time, outputted to the second coefficient calculatingunit 1750 in order to update the coefficient.

The CIR estimator 1741 of the first coefficient calculating unit 1740uses the data received during the known data section and the known datain order to estimate the CIR. Subsequently, the estimated CIR isoutputted to the interpolator 1742. The interpolator 1742 uses theinputted CIR to estimate the CIRs (i.e., CIRs of the region that doesnot include the known data) corresponding to the points located betweenthe estimated CIRs according to a predetermined interpolation method.Thereafter, the estimated result is outputted to the second FFT unit1743. The second FFT unit 1743 performs FFT on the inputted CIR, so asto convert the inputted CIR to a frequency domain CIR. Thereafter, theconverted frequency domain CIR is outputted to the coefficientcalculator 1744. The coefficient calculator 1744 calculates a frequencydomain equalization coefficient that satisfies the condition of usingthe CIR of the frequency domain so as to minimize the mean square error.The calculated equalizer coefficient of the frequency domain is thenoutputted to the coefficient calculator 1760.

The structure and operations of the second coefficient calculating unit1750 is identical to those of the second coefficient calculating unit1650 shown in FIG. 38. Therefore, the description of the same will beomitted for simplicity. If the input data correspond to the data ofregion A, the coefficient selector 1760 selects the equalizationcoefficient calculated from the first coefficient calculating unit 1740.On the other hand, if the input data correspond to the data of regions Band C, the coefficient selector 1760 selects the equalizationcoefficient updated by the second coefficient calculating unit 1750.Thereafter, the selected equalization coefficient is outputted to thedistortion compensator 1720.

FIG. 40 illustrates a block diagram of a channel equalizer according toanother embodiment of the present invention. In other words, FIG. 40illustrates a block diagram showing another example of a channelequalizer by using different CIR estimation and application methods inaccordance with regions A, B, and C, when the data group is divided intothe structure shown in FIG. 6A. For example, in region A, the presentinvention uses the known data in order to estimate the CIR by using aleast square (LS) method, thereby performing the channel equalizationprocess. On the other hand, in regions B and C, the present inventionestimates the CIR by using a least mean square (LMS) method, therebyperforming the channel equalization process. More specifically, sincethe periodic known data do not exist in regions B and C, as in region A,the same channel equalization process as that of region A cannot beperformed in regions B and C. Therefore, the channel equalizationprocess may only be performed by using the LMS method.

Referring to FIG. 40, the channel equalizer includes an overlap unit1801, a first fast fourier transform (FFT) unit 1802, a distortioncompensator 1803, an inverse fast fourier transform (IFFT) unit 1804, asave unit 1805, a first CIR estimator 1806, a CIR interpolator 1807, adecision unit 1808, a second CIR estimator 1810, a selection unit 1811,a second FFT unit 1812, and a coefficient calculator 1813. Herein, anydevice performed complex number multiplication may be used as thedistortion compensator 1803. In the channel equalizer having theabove-described structure, as shown in FIG. 40, the overlap unit 1801overlaps the data being inputted to the channel equalizer to apredetermined overlapping ratio and then outputs the overlapped data tothe first FFT unit 1802. The first FFT unit 1802 converts (ortransforms) the overlapped data of the time domain to overlapped data ofthe frequency domain by using fast fourier transform (FFT). Then, theconverted data are outputted to the distortion compensator 1803.

The distortion converter 1803 performs complex multiplication on theequalization coefficient calculated from the coefficient calculator 1813and the overlapped data of the frequency domain, thereby compensatingthe channel distortion of the overlapped data being outputted from thefirst FFT unit 1802. Thereafter, the distortion-compensated data areoutputted to the IFFT unit 1804. The IFFT unit 1804 performs inversefast fourier transform (IFFT) on the distortion-compensated overlappeddata, so as to convert the corresponding data back to data (i.e.,overlapped data) of the time domain. Subsequently, the converted dataare outputted to the save unit 1805. The save unit 1805 extracts onlythe valid data from the overlapped data of the time domain. Then, thesave unit 1805 outputs the extracted valid data for a data decodingprocess and, at the same time, outputs the extracted valid data to thedecision unit 1808 for a channel estimation process.

The decision unit 1808 selects one of a plurality of decision values(e.g., 8 decision values) that is most approximate to the equalized dataand outputs the selected decision value to the select unit 1809. Herein,a multiplexer may be used as the select unit 1809. In a general datasection, the select unit 1809 selects the decision value of the decisionunit 1808. Alternatively, in a known data section, the select unit 1809selects the known data and outputs the selected known data to the secondCIR estimator 1810. Meanwhile, the first CIR estimator 1806 uses thedata that are being inputted in the known data section and the knowndata so as to estimate the CIR.

Thereafter, the first CIR estimator 1806 outputs the estimated CIR tothe CIR interpolator 1807. Herein, the known data correspond toreference known data created during the known data section by thereceiving system in accordance to an agreement between the transmittingsystem and the receiving system. At this point, according to anembodiment of the present invention, the first CIR estimator 1806 usesthe LS method to estimate the CIR. The LS estimation method calculates across correlation value p between the known data that have passedthrough the channel during the known data section and the known datathat are already known by the receiving end. Then, a cross correlationmatrix R of the known data is calculated. Subsequently, a matrixoperation is performed on R⁻¹·p so that the cross correlation portionwithin the cross correlation value p between the received data and theinitial known data, thereby estimating the CIR of the transmissionchannel.

The CIR interpolator 1807 receives the CIR from the first CIR estimator1806. And, in the section between two sets of known data, the CIR isinterpolated in accordance with a pre-determined interpolation method.Then, the interpolated CIR is outputted. At this point, thepre-determined interpolation method corresponds to a method ofestimating a particular set of data at an unknown point by using a setof data known by a particular function. For example, such methodincludes a linear interpolation method. The linear interpolation methodis only one of the most simple interpolation methods. A variety of otherinterpolation methods may be used instead of the above-described linearinterpolation method. It is apparent that the present invention is notlimited only to the example set forth in the description of the presentinvention. More specifically, the CIR interpolator 1807 uses the CIRthat is being inputted in order to estimate the CIR of the section thatdoes not include any known data by using the pre-determinedinterpolation method. Thereafter, the estimated CIR is outputted to theselect unit 1811.

The second CIR estimator 1810 uses the input data of the channelequalizer and the output data of the select unit 1809 in order toestimate the CIR. Then, the second CIR estimator 1810 outputs theestimated CIR to the select unit 1811. At this point, according to anembodiment of the present invention, the CIR is estimated by using theLMS method. The LMS estimation method will be described in detail in alater process. In region A, the select unit 1811 selects the CIRoutputted from the CIR interpolator 1807. And, in regions B and C, theselect unit 1811 selects the CIR outputted from the second CIR estimator1810. Thereafter, the select unit 1811 outputs the selected CIR to thesecond FFT unit 1812.

The second FFT unit 1812 converts the CIR that is being inputted to aCIR of the frequency domain, which is then outputted to the coefficientcalculator 1813. The coefficient calculator 1813 uses the CIR of thefrequency domain that is being inputted, so as to calculate theequalization coefficient and to output the calculated equalizationcoefficient to the distortion compensator 1803. At this point, thecoefficient calculator 1813 calculates a channel equalizationcoefficient of the frequency domain that can provide minimum mean squareerror (MMSE) from the CIR of the frequency domain. At this point, thesecond CIR estimator 1810 may use the CIR estimated in region A as theCIR at the beginning of regions B and C. For example, the last CIR valueof region A may be used as the CIR value at the beginning of the Cregion. Accordingly, the convergence speed of regions B and C may bereduced.

The basic principle of estimating the CIR by using the LMS method in thesecond CIR estimator 1810 corresponds to receiving the output of anunknown transmission channel and to updating (or renewing) thecoefficient of an adaptive filter (not shown) so that the differencevalue between the output value of the unknown channel and the outputvalue of the adaptive filter is minimized. More specifically, thecoefficient value of the adaptive filter is renewed so that the inputdata of the channel equalizer is equal to the output value of theadaptive filter (not shown) included in the second CIR estimator 1810.Thereafter, the filter coefficient is outputted as the CIR after eachFFT cycle.

Referring to FIG. 41, the second CIR estimator 1810 includes a delayunit T, a multiplier, and a coefficient renewal unit for each tab.Herein, the delay unit T sequentially delays the output data {circumflexover (x)}(n) of the select unit 1809. The multiplier multipliesrespective output data outputted from each delay unit T with error datae(n) The coefficient renewal unit renews the coefficient by using theoutput corresponding to each multiplier. Herein, the multipliers thatare being provided as many as the number of tabs will be referred to asa first multiplying unit for simplicity. Furthermore, the second CIRestimator 1810 further includes a plurality of multipliers eachmultiplying the output data of the select unit 1809 and the output dataof the delay unit T (wherein the output data of the last delay unit areexcluded) with the output data corresponding to each respectivecoefficient renewal unit. These multipliers are also provided as many asthe number of tabs. This group of multipliers will be referred to as asecond multiplying unit for simplicity.

The second CIR estimator 1810 further includes an adder and asubtractor. Herein, the adder adds all of the data outputted from eachmultipliers included in the second multiplier unit. Then, the addedvalue is outputted as the estimation value ŷ(n) of the data inputted tothe channel equalizer. The subtractor calculates the difference betweenthe output data ŷ(n) of the adder and the input data y(n) of the channelequalizer. Thereafter, the calculated difference value is outputted asthe error data e(n). Referring to FIG. 41, in a general data section,the decision value of the equalized data is inputted to the first delayunit included in the second CIR estimator 1810 and to the firstmultiplier included in the second multiplier. In the known data section,the known data are inputted to the first delay unit included in thesecond CIR estimator 1810 and to the first multiplier included in thesecond multiplier unit. The input data {circumflex over (x)}(n) aresequentially delayed by passing through a number of serially connecteddelay units T, the number corresponding to the number of tabs. Theoutput data of each delay unit T and the error data e(n) are multipliedby each corresponding multiplier included in the first multiplier unit.Thereafter, the coefficients are renewed by each respective coefficientrenewal unit.

Each coefficient that is renewed by the corresponding coefficientrenewal unit is multiplied with the input data the output data{circumflex over (x)}(n) and also with the output data of each delayunit T with the exception of the last delay. Thereafter, the multipliedvalue is inputted to the adder. The adder then adds all of the outputdata outputted from the second multiplier unit and outputs the addedvalue to the subtractor as the estimation value ŷ(n) of the input dataof the channel equalizer. The subtractor calculates a difference valuebetween the estimation value ŷ(n) and the input data y(n) of the channelequalizer. The difference value is then outputted to each multiplier ofthe first multiplier unit as the error data e(n). At this point, theerror data e(n) is outputted to each multiplier of the first multiplierunit by passing through each respective delay unit T. As describedabove, the coefficient of the adaptive filter is continuously renewed.And, the output of each coefficient renewal unit is outputted as the CIRof the second CIR estimator 1810 after each FFT cycle.

FIG. 42 illustrates a block diagram of a channel equalizer according toanother embodiment of the present invention. Herein, by estimating andcompensating a remaining carrier phase error from a channel-equalizedsignal, the receiving system of the present invention may be enhanced.Referring to FIG. 42, the channel equalizer includes a first frequencydomain converter 2100, a channel estimator 2110, a second frequencydomain converter 2121, a coefficient calculator 2122, a distortioncompensator 2130, a time domain converter 2140, a remaining carrierphase error remover 2150, a noise canceller (NC) 2160, and a decisionunit 2170.

Herein, the first frequency domain converter 2100 includes an overlapunit 2101 overlapping inputted data, and a fast fourier transform (FFT)unit 2102 converting the data outputted from the overlap unit 2101 tofrequency domain data. The channel estimator 2110 includes a CIRestimator 2111 estimating a CIR from the inputted data, a phasecompensator 2112 compensating the phase of the CIR estimated by the CIRestimator 2111, and a linear interpolator 2113 performing linearinterpolation on the CIR having its phase compensated. The secondfrequency domain converter 2121 includes a FFT unit converting the CIRbeing outputted from the channel estimator 2110 to a frequency domainCIR.

The time domain converter 2140 includes an IFFT unit 2141 converting thedata having the distortion compensated by the distortion compensator2130 to time domain data, and a save unit 2142 extracting only validdata from the data outputted from the IFFT unit 2141. The remainingcarrier phase error remover 2150 includes an error compensator 2151removing the remaining carrier phase error included in the channelequalized data, and a remaining carrier phase error estimator 2152 usingthe channel equalized data and the decision data of the decision unit2170 so as to estimate the remaining carrier phase error, therebyoutputting the estimated error to the error compensator 2151. Herein,any device performing complex number multiplication may be used as thedistortion compensator 2130 and the error compensator 2151.

At this point, since the received data correspond to data modulated toVSB type data, 8-level scattered data exist only in the real numberelement. Therefore, referring to FIG. 42, all of the signals used in thenoise canceller 2160 and the decision unit 2170 correspond to realnumber (or in-phase) signals. However, in order to estimate andcompensate the remaining carrier phase error and the phase noise, bothreal number (in-phase) element and imaginary number (quadrature) elementare required. Therefore, the remaining carrier phase error remover 2150receives and uses the quadrature element as well as the in-phaseelement. Generally, prior to performing the channel equalizationprocess, the demodulator 902 within the receiving system performsfrequency and phase recovery of the carrier. However, if a remainingcarrier phase error that is not sufficiently compensated is inputted tothe channel equalizer, the performance of the channel equalizer may bedeteriorated. Particularly, in a dynamic channel environment, theremaining carrier phase error may be larger than in a static channelenvironment due to the frequent and sudden channel changes. Eventually,this acts as an important factor that deteriorates the receivingperformance of the present invention.

Furthermore, a local oscillator (not shown) included in the receivingsystem should preferably include a single frequency element. However,the local oscillator actually includes the desired frequency elements aswell as other frequency elements. Such unwanted (or undesired) frequencyelements are referred to as phase noise of the local oscillator. Suchphase noise also deteriorates the receiving performance of the presentinvention. It is difficult to compensate such remaining carrier phaseerror and phase noise from the general channel equalizer. Therefore, thepresent invention may enhance the channel equaling performance byincluding a carrier recovery loop (i.e., a remaining carrier phase errorremover 2150) in the channel equalizer, as shown in FIG. 42, in order toremove the remaining carrier phase error and the phase noise.

More specifically, the receiving data demodulated in FIG. 42 areoverlapped by the overlap unit 2101 of the first frequency domainconverter 2100 at a pre-determined overlapping ratio, which are thenoutputted to the FFT unit 2102. The FFT unit 2102 converts theoverlapped time domain data to overlapped frequency domain data throughby processing the data with FFT. Then, the converted data are outputtedto the distortion compensator 2130. The distortion compensator 2130performs a complex number multiplication on the overlapped frequencydomain data outputted from the FFT unit 2102 included in the firstfrequency domain converter 2100 and the equalization coefficientcalculated from the coefficient calculator 2122, thereby compensatingthe channel distortion of the overlapped data outputted from the FFTunit 2102. Thereafter, the compensated data are outputted to the IFFTunit 2141 of the time domain converter 2140. The IFFT unit 2141 performsIFFT on the overlapped data having the channel distortion compensated,thereby converting the overlapped data to time domain data, which arethen outputted to the error compensator 2151 of the remaining carrierphase error remover 2150.

The error compensator 2151 multiplies a signal compensating theestimated remaining carrier phase error and phase noise with the validdata extracted from the time domain. Thus, the error compensator 2151removes the remaining carrier phase error and phase noise included inthe valid data. The data having the remaining carrier phase errorcompensated by the error compensator 2151 are outputted to the remainingcarrier phase error estimator 2152 in order to estimate the remainingcarrier phase error and phase noise and, at the same time, outputted tothe noise canceller 2160 in order to remove (or cancel) the noise.

The remaining carrier phase error estimator 2152 uses the output data ofthe error compensator 2151 and the decision data of the decision unit2170 to estimate the remaining carrier phase error and phase noise.Thereafter, the remaining carrier phase error estimator 2152 outputs asignal for compensating the estimated remaining carrier phase error andphase noise to the error compensator 2151. In this embodiment of thepresent invention, an inverse number of the estimated remaining carrierphase error and phase noise is outputted as the signal for compensatingthe remaining carrier phase error and phase noise.

FIG. 43 illustrates a detailed block diagram of the remaining carrierphase error estimator 2152 according to an embodiment of the presentinvention. Herein, the remaining carrier phase error estimator 2152includes a phase error detector 2211, a loop filter 2212, a numericallycontrolled oscillator (NCO) 2213, and a conjugator 2214. Referring toFIG. 43, the decision data, the output of the phase error detector 2211,and the output of the loop filter 2212 are all real number signals. And,the output of the error compensator 2151, the output of the NCO 2213,and the output of the conjugator 2214 are all complex number signals.

The phase error detector 2211 receives the output data of the errorcompensator 2151 and the decision data of the decision unit 2170 inorder to estimate the remaining carrier phase error and phase noise.Then, the phase error detector 2211 outputs the estimated remainingcarrier phase error and phase noise to the loop filter. The loop filter2212 then filters the remaining carrier phase error and phase noise,thereby outputting the filtered result to the NCO 2213. The NCO 2213generates a cosine corresponding to the filtered remaining carrier phaseerror and phase noise, which is then outputted to the conjugator 2214.

The conjugator 2214 calculates the conjugate value of the cosine wavegenerated by the NCO 2213. Thereafter, the calculated conjugate value isoutputted to the error compensator 2151. At this point, the output dataof the conjugator 2214 becomes the inverse number of the signalcompensating the remaining carrier phase error and phase noise. In otherwords, the output data of the conjugator 2214 becomes the inverse numberof the remaining carrier phase error and phase noise.

The error compensator 2151 performs complex number multiplication on theequalized data outputted from the time domain converter 2140 and thesignal outputted from the conjugator 2214 and compensating the remainingcarrier phase error and phase noise, thereby removing the remainingcarrier phase error and phase noise included in the equalized data.Meanwhile, the phase error detector 2211 may estimate the remainingcarrier phase error and phase noise by using diverse methods andstructures. According to this embodiment of the present invention, theremaining carrier phase error and phase noise are estimated by using adecision-directed method.

If the remaining carrier phase error and phase noise are not included inthe channel-equalized data, the decision-directed phase error detectoraccording to the present invention uses the fact that only real numbervalues exist in the correlation values between the channel-equalizeddata and the decision data. More specifically, if the remaining carrierphase error and phase noise are not included, and when the input data ofthe phase error detector 2211 are referred to as x_(i)+jx_(q), thecorrelation value between the input data of the phase error detector2211 and the decision data may be obtained by using Equation 8 shownbelow:

E{(x _(i) +jx _(q))({circumflex over (x)} _(i) +j{circumflex over (x)}_(q))*}  Equation 8

At this point, there is no correlation between x_(i) and x_(q).Therefore, the correlation value between x_(i) and x_(q) is equal to 0.Accordingly, if the remaining carrier phase error and phase noise arenot included, only the real number values exist herein. However, if theremaining carrier phase error and phase noise are included, the realnumber element is shown in the imaginary number value, and the imaginarynumber element is shown in the real number value. Thus, in this case,the imaginary number element is shown in the correlation value.Therefore, it can be assumed that the imaginary number portion of thecorrelation value is in proportion with the remaining carrier phaseerror and phase noise. Accordingly, as shown in Equation 9 below, theimaginary number of the correlation value may be used as the remainingcarrier phase error and phase noise.

Phase Error=imag{(x _(i) +jx _(q))({circumflex over (x)} _(i)+j{circumflex over (x)} _(q))*}

Phase Error=x _(q) {circumflex over (x)} _(i) −x _(i) {circumflex over(x)} _(q)  Equation 9

FIG. 44 illustrates a block diagram of a phase error detector 2211obtaining the remaining carrier phase error and phase noise. Herein, thephase error detector 2211 includes a Hilbert converter 2311, a complexnumber configurator 2312, a conjugator 2313, a multiplier 2314, and aphase error output 2315. More specifically, the Hilbert converter 2311creates an imaginary number decision data {circumflex over (x)}_(q) byperforming a Hilbert conversion on the decision value {circumflex over(x)}_(i) of the decision unit 2170. The generated imaginary numberdecision value is then outputted to the complex number configurator2312. The complex number configurator 2312 uses the decision data{circumflex over (x)}_(i) and {circumflex over (x)}_(q) to configure thecomplex number decision data {circumflex over (x)}_(i)+j{circumflex over(x)}_(q), which are then outputted to the conjugator 2313. Theconjugator 2313 conjugates the output of the complex number configurator2312, thereby outputting the conjugated value to the multiplier 2314.The multiplier 2314 performs a complex number multiplication on theoutput data of the error compensator 2151 and the output data{circumflex over (x)}_(i)−j{circumflex over (x)}_(q) of the conjugator2313, thereby obtaining the correlation between the output datax_(i)+jx_(q) of the error compensator 2151 and the decision value{circumflex over (x)}_(i)−j{circumflex over (x)}_(q) of the decisionunit 2170. The correlation data obtained from the multiplier 2314 arethen inputted to the phase error output 2315. The phase error output2315 outputs the imaginary number portion x_(q){circumflex over(x)}_(i)−x_(i){circumflex over (x)}_(q) of the correlation dataoutputted from the multiplier 2314 as the remaining carrier phase errorand phase noise.

The phase error detector shown in FIG. 44 is an example of a pluralityof phase error detecting methods. Therefore, other types of phase errordetectors may be used in the present invention. Therefore, the presentinvention is not limited only to the examples and embodiments presentedin the description of the present invention. Furthermore, according toanother embodiment of the present invention, at least 2 phase errordetectors are combined so as to detect the remaining carrier phase errorand phase noise. Accordingly, the output of the remaining carrier phaseerror remover 2150 having the detected remaining carrier phase error andphase noise removed as described above, is configured of an addition ofthe original (or initial) signal having the channel equalization, theremaining carrier phase error and phase noise, and the signalcorresponding to a white noise being amplified to a colored noise duringthe channel equalization.

Therefore, the noise canceller 2160 receives the output data of theremaining carrier phase error remover 2150 and the decision data of thedecision unit 2170, thereby estimating the colored noise. Then, thenoise canceller 2160 subtracts the estimated colored noise from the datahaving the remaining carrier phase error and phase noise removedtherefrom, thereby removing the noise amplified during the equalizationprocess. The data having the noise removed (or cancelled) by the noisecanceller 2160 are outputted for the data decoding process and, at thesame time, outputted to the decision unit 2170.

The decision unit 2170 selects one of a plurality of pre-determineddecision data sets (e.g., 8 decision data sets) that is most approximateto the output data of the noise canceller 2160, thereby outputting theselected data to the remaining carrier phase error estimator 2152 andthe noise canceller 2160. Meanwhile, the received data are inputted tothe overlap unit 2101 of the first frequency domain converter 2100included in the channel equalizer and, at the same time, inputted to theCIR estimator 2111 of the channel estimator 2110. The CIR estimator 2111uses a training sequence, for example, data being inputted during theknown data section and the known data in order to estimate the CIR,thereby outputting the estimated CIR to the phase compensator 2112.Herein, the known data correspond to reference known data generatedduring the known data section by the receiving system in accordance withan agreement between the receiving system and the transmitting system.

Furthermore, in this embodiment of the present invention, the CIRestimator 2111 estimates the CIR by using the least square (LS) method.The LS estimation method calculates a cross correlation value p betweenthe known data that have passed through the channel during the knowndata section and the known data that are already known by the receivingend. Then, a cross correlation matrix R of the known data is calculated.Subsequently, a matrix operation is performed on R⁻¹·p so that the crosscorrelation portion within the cross correlation value p between thereceived data and the initial known data, thereby estimating the CIR ofthe transmission channel.

The phase compensator 2112 compensates the phase change of the estimatedCIR. Then, the phase compensator 2112 outputs the compensated CIR to thelinear interpolator 2113. At this point, the phase compensator 2112 maycompensate the phase change of the estimated CIR by using a maximumlikelihood method. More specifically, the remaining carrier phase errorand phase noise that are included in the demodulated received data and,therefore, being inputted change the phase of the CIR estimated by theCIR estimator 2111 at a cycle period of one known data sequence. At thispoint, if the phase change of the inputted CIR, which is to be used forthe linear interpolation process, is not performed in a linear form dueto a high rate of the phase change, the channel equalizing performanceof the present invention may be deteriorated when the channel iscompensated by calculating the equalization coefficient from the CIR,which is estimated by a linear interpolation method.

Therefore, the present invention removes (or cancels) the amount ofphase change of the CIR estimated by the CIR estimator 2111 so that thedistortion compensator 2130 allows the remaining carrier phase error andphase noise to bypass the distortion compensator 2130 without beingcompensated. Accordingly, the remaining carrier phase error and phasenoise are compensated by the remaining carrier phase error remover 2150.For this, the present invention removes (or cancels) the amount of phasechange of the CIR estimated by the phase compensator 2112 by using amaximum likelihood method. The basic idea of the maximum likelihoodmethod relates to estimating a phase element mutually (or commonly)existing in all CIR elements, then to multiply the estimated CIR with aninverse number of the mutual (or common) phase element, so that thechannel equalizer, and most particularly, the distortion compensator2130 does not compensate the mutual phase element.

More specifically, when the mutual phase element is referred to as θ,the phase of the newly estimated CIR is rotated by θ as compared to thepreviously estimated CIR. When the CIR of a point t is referred to ash_(i)(t), the maximum likelihood phase compensation method obtains aphase θ_(ML) corresponding to when h_(i)(t) is rotated by θ, the squaredvalue of the difference between the CIR of h_(i)(t) and the CIR ofh_(i)(t+1), i.e., the CIR of a point (t+1), becomes a minimum value.Herein, when i represents a tap of the estimated CIR, and when Nrepresents a number of taps of the CIR being estimated by the CIRestimator 2111, the value of θ_(ML) is equal to or greater than 0 andequal to or less than N−1. This value may be calculated by usingEquation 10 shown below:

$\begin{matrix}{\theta_{ML} = {\min\limits_{\theta}{\sum\limits_{i = 0}^{N - 1}{{{{h_{i}(t)}^{j\theta}} - {h_{i}\left( {t + 1} \right)}}}^{2}}}} & {{Equation}\mspace{14mu} 10}\end{matrix}$

Herein, in light of the maximum likelihood method, the mutual phaseelement θ_(ML) is equal to the value of θ, when the right side ofEquation 10 being differentiated with respect to θ is equal to 0. Theabove-described condition is shown in Equation 11 below:

$\begin{matrix}{{\frac{}{\theta}{\sum\limits_{i = 0}^{N - 1}{{{{h_{i}(t)}^{j\theta}} - {h_{i}\left( {t + 1} \right)}}}^{2}}} = {{\frac{}{\theta}{\sum\limits_{i = 0}^{N - 1}{\begin{pmatrix}{{{h_{i}(t)}^{j\theta}} -} \\{h_{i}\left( {t + 1} \right)}\end{pmatrix}\begin{pmatrix}{{{h_{i}(t)}^{j\theta}} -} \\{h_{i}\left( {t + 1} \right)}\end{pmatrix}^{*}}}} = {{\frac{}{\theta}{\sum\limits_{i = 0}^{N - 1}\begin{Bmatrix}{{{h_{i}(t)}}^{2} + {{h_{i + 1}(t)}}^{2} -} \\{{{h_{i}(t)}{h_{i}^{*}\left( {t + 1} \right)}^{j\theta}} -} \\{{h_{i}^{*}(t)}{h_{i}\left( {t + 1} \right)}^{- {j\theta}}}\end{Bmatrix}}} = {{\sum\limits_{i = 0}^{N - 1}\begin{Bmatrix}{{j\; {h_{i}^{*}(t)}{h_{i}\left( {t + 1} \right)}^{- {j\theta}}} -} \\{j\; {h_{i}^{*}(t)}\left( {t + 1} \right)^{j\theta}}\end{Bmatrix}} = {{j{\sum\limits_{i = 0}^{N - 1}{2{Im}\left\{ {{h_{i}^{*}(t)}{h_{i}\left( {t + 1} \right)}^{- {j\theta}}} \right\}}}} = 0}}}}} & {{Equation}\mspace{14mu} 11}\end{matrix}$

The above Equation 11 may be simplified as shown in Equation 12 below:

$\begin{matrix}{{{{Im}\left\{ {^{- {j\theta}}{\sum\limits_{i = 0}^{N - 1}\left\{ {{h_{i}^{*}(t)}{h_{i}\left( {t + 1} \right)}} \right\}}} \right\}} = 0}{\theta_{ML} = {\arg \left( {\sum\limits_{i = 0}^{N - 1}{{h_{i}^{*}(t)}{h_{i}\left( {t + 1} \right)}}} \right)}}} & {{Equation}\mspace{14mu} 12}\end{matrix}$

More specifically, Equation 12 corresponds to the θ_(ML) value that isto be estimated by the argument of the correlation value between h_(i)(t) and h_(i)(t+1).

FIG. 45 illustrates a phase compensator according to an embodiment ofthe present invention, wherein the mutual phase element θ_(ML) iscalculated as described above, and wherein the estimated phase elementis compensated at the estimated CIR. Referring to FIG. 45, the phasecompensator includes a correlation calculator 2410, a phase changeestimator 2420, a compensation signal generator 2430, and a multiplier2440. The correlation calculator 2410 includes a first N symbol buffer2411, an N symbol delay 2412, a second N symbol buffer 2413, aconjugator 2414, and a multiplier 2415. More specifically, the first Nsymbol buffer 2411 included in the correlation calculator 2410 iscapable of storing the data being inputted from the CIR estimator 2111in symbol units to a maximum limit of N number of symbols. The symboldata being temporarily stored in the first N symbol buffer 2411 are theninputted to the multiplier 2415 included in the correlation calculator2410 and to the multiplier 2440.

At the same time, the symbol data being outputted from the CIR estimator2111 are delayed by N symbols from the N symbol delay 2412. Then, thedelayed symbol data pass through the second N symbol buffer 2413 andinputted to the conjugator 2414, so as to be conjugated and theninputted to the multiplier 2415. The multiplier 2415 multiplies theoutput of the first N symbol buffer 2411 and the output of theconjugator 2414. Then, the multiplier 2415 outputs the multiplied resultto an accumulator 2421 included in the phase change estimator 2420. Morespecifically, the correlation calculator 2410 calculates a correlationbetween a current CIR h_(i)(t+1) having the length of N and a previousCIR h_(i)(t) also having the length of N. then, the correlationcalculator 2410 outputs the calculated correlation value to theaccumulator 2421 of the phase change estimator 2420.

The accumulator 2421 accumulates the correlation values outputted fromthe multiplier 2415 during an N symbol period. Then, the accumulator2421 outputs the accumulated value to the phase detector 2422. The phasedetector 2422 then calculates a mutual phase element θ_(ML) from theoutput of the accumulator 2421 as shown in the above-described Equation11. Thereafter, the calculated θ_(ML) value is outputted to thecompensation signal generator 2430. The compensation signal generator2430 outputs a complex signal e^(−jθ) ^(ML) having a phase opposite tothat of the detected phase as the phase compensation signal to themultiplier 2440. The multiplier 2440 multiplies the current CIRh_(i)(t+1) being outputted from the first N symbol buffer 2411 with thephase compensation signal e^(−jθ) ^(ML) , thereby removing the amount ofphase change of the estimated CIR.

As described above, the phase compensator 2112 using the maximumlikelihood method calculates a phase element corresponding to thecorrelation value between the inputted CIR and the previous CIR beingdelayed by N symbols. Thereafter, the phase compensator 2112 generates aphase compensation signal having a phase opposite to that of the phaseelement calculated as described above. Subsequently, the phasecompensator 2112 multiplies the generated phase compensation signal tothe estimated CIR, thereby removing the amount of phase change of theestimated CIR. The CIR having the phase change compensated is inputtedto the linear interpolator 2113. The linear interpolator 2113 linearlyinterpolates the CIRs having the phase changes compensated to the secondfrequency domain converter 2121.

More specifically, the linear interpolator 2113 receives the CIR havingthe phase change compensated from the phase compensator 2112.Accordingly, during the known data section, the linear interpolator 2113outputs the received CIR, and during the data section in-between knowndata, the CIR is interpolated in accordance with a pre-determinedinterpolating method. Thereafter, the interpolated CIR is outputted. Inthis embodiment of the present invention, a linear interpolation method,which is one of the many pre-determined interpolating methods, is usedto interpolate the CIRs. Herein, the present invention may also useother interpolation methods. Therefore, the present invention is notlimited only to the examples given in the description of the presentinvention.

The second frequency domain converter 2121 performs FFT on the CIR beingoutputted from the linear interpolator 2113, thereby converting the CIRto a frequency domain CIR. Then, the second frequency domain converter2121 outputs the converted CIR to the coefficient calculator 2122. Thecoefficient calculator 2122 uses the frequency domain CIR beingoutputted from the second frequency domain converter 2121 to calculatethe equalization coefficient. Then, the coefficient calculator 2122outputs the calculated coefficient to the distortion compensator 2130.Herein, for example, the coefficient calculator 2122 calculates achannel equalization coefficient of the frequency domain that canprovide minimum mean square error (MMSE) from the CIR of the frequencydomain, which is outputted to the distortion compensator 2130. Thedistortion compensator 2130 performs a complex number multiplication onthe overlapped data of the frequency domain being outputted from the FFTunit 2102 of the first frequency domain converter 2100 and theequalization coefficient calculated by the coefficient calculator 2122,thereby compensating the channel distortion of the overlapped data beingoutputted from the FFT unit 2102.

Meanwhile, if the data being inputted to the block decoder 905 afterbeing channel equalized from the equalizer 903 correspond to the mobileservice data having additional encoding and trellis encoding performedthereon by the transmitting system, trellis decoding and additionaldecoding processes are performed on the inputted data as inverseprocesses of the transmitting system. Alternatively, if the data beinginputted to the block decoder 905 correspond to the main service datahaving only trellis encoding performed thereon, and not the additionalencoding, only the trellis decoding process is performed on the inputteddata as the inverse process of the transmitting system. The data groupdecoded by the block decoder 905 is inputted to the data deformatter906, and the main service data packet is inputted to the datadeinterleaver 909.

More specifically, if the inputted data correspond to the main servicedata, the block decoder 905 performs Viterbi decoding on the inputteddata so as to output a hard decision value or to perform a hard-decisionon a soft decision value, thereby outputting the result. Meanwhile, ifthe inputted data correspond to the mobile service data, the blockdecoder 905 outputs a hard decision value or a soft decision value withrespect to the inputted mobile service data. In other words, if theinputted data correspond to the mobile service data, the block decoder905 performs a decoding process on the data encoded by the blockprocessor and trellis encoding module of the transmitting system.

At this point, the RS frame encoder of the pre-processor included in thetransmitting system may be viewed as an external code. And, the blockprocessor and the trellis encoder may be viewed as an internal code. Inorder to maximize the performance of the external code when decodingsuch concatenated codes, the decoder of the internal code should outputa soft decision value. Therefore, the block decoder 905 may output ahard decision value on the mobile service data. However, when required,it may be more preferable for the block decoder 905 to output a softdecision value.

Meanwhile, the data deinterleaver 909, the RS decoder 910, and thederandomizer 911 are blocks required for receiving the main servicedata. Therefore, the above-mentioned blocks may not be required in thestructure of a digital broadcast receiving system that only receives themobile service data. The data deinterleaver 909 performs an inverseprocess of the data interleaver included in the transmitting system. Inother words, the data deinterleaver 909 deinterleaves the main servicedata outputted from the block decoder 905 and outputs the deinterleavedmain service data to the RS decoder 910. The RS decoder 910 performs asystematic RS decoding process on the deinterleaved data and outputs theprocessed data to the derandomizer 911. The derandomizer 911 receivesthe output of the RS decoder 910 and generates a pseudo random data byteidentical to that of the randomizer included in the digital broadcasttransmitting system. Thereafter, the derandomizer 911 performs a bitwiseexclusive OR (XOR) operation on the generated pseudo random data byte,thereby inserting the MPEG synchronization bytes to the beginning ofeach packet so as to output the data in 188-byte main service datapacket units.

Meanwhile, the data being outputted from the block decoder 905 to thedata deformatter 906 are inputted in the form of a data group. At thispoint, the data deformatter 906 already knows the structure of the datathat are to be inputted and is, therefore, capable of identifying thesignaling information, which includes the system information, and themobile service data from the data group. Thereafter, the datadeformatter 906 outputs the identified signaling information to a blockfor system information and outputs the identified mobile service data tothe RS frame decoder 907. At this point, the data deformatter 906removes the known data, trellis initialization data, and MPEG headerthat were inserted in the main service data and data group. The datadeformatter 906 also removes the RS parity that was added by the RSencoder/non-systematic RS encoder or the non-systematic RS encoder ofthe transmitting system. Thereafter, the processed data are outputted tothe RS frame decoder 907. More specifically, the RS frame decoder 907receives only the RS encoded and CRC encoded mobile service data thatare transmitted from the data deformatter 906.

The RS frame encoder 907 performs an inverse process of the RS frameencoder included in the transmitting system so as to correct the errorwithin the RS frame. Then, the RS frame decoder 907 adds the 1-byte MPEGsynchronization service data packet, which had been removed during theRS frame encoding process, to the error-corrected mobile service datapacket. Thereafter, the processed data packet is outputted to thederandomizer 908. The operation of the RS frame decoder 907 will bedescribed in detail in a later process. The derandomizer 908 performs aderandomizing process, which corresponds to the inverse process of therandomizer included in the transmitting system, on the received mobileservice data. Thereafter, the derandomized data are outputted, therebyobtaining the mobile service data transmitted from the transmittingsystem.

FIG. 46 illustrates a series of exemplary step of an error correctiondecoding process of the RS frame decoder 907 according to the presentinvention. More specifically, the RS frame decoder 907 groups mobileservice data bytes received from the data deformatter 906 so as toconfigure an RS frame. The mobile service data correspond to data RSencoded and CRC encoded from the transmitting system. FIG. 46( a)illustrates an example of configuring the RS frame. More specifically,the transmitting system divided the RS frame having the size of (N+2)235to 30*235 byte blocks. When it is assumed that each of the dividedmobile service data byte blocks is inserted in each data group and thentransmitted, the receiving system also groups the 30*235 mobile servicedata byte blocks respectively inserted in each data group, therebyconfiguring an RS frame having the size of (N+2)235. For example, whenit is assumed that an RS frame is divided into 18 30*235 byte blocks andtransmitted from a burst section, the receiving system also groups themobile service data bytes of 18 data groups within the correspondingburst section, so as to configure the RS frame. Furthermore, when it isassumed that N is equal to 538 (i.e., N=538), the RS frame decoder 907may group the mobile service data bytes within the 18 data groupsincluded in a burst so as to configure a RS frame having the size of540*235 bytes.

Herein, when it is assumed that the block decoder 905 outputs a softdecision value for the decoding result, the RS frame decoder 907 maydecide the ‘0’ and ‘1’ of the corresponding bit by using the codes ofthe soft decision value. 8 bits that are each decided as described aboveare grouped to create 1 data byte. If the above-described process isperformed on all soft decision values of the 18 data groups included ina single burst, the RS frame having the size of 540*235 bytes may beconfigured. Additionally, the present invention uses the soft decisionvalue not only to configure the RS frame but also to configure areliability map. Herein, the reliability map indicates the reliabilityof the corresponding data byte, which is configured by grouping 8 bits,the 8 bits being decided by the codes of the soft decision value.

For example, when the absolute value of the soft decision value exceedsa pre-determined threshold value, the value of the corresponding bit,which is decided by the code of the corresponding soft decision value,is determined to be reliable. Conversely, when the absolute value of thesoft decision value does not exceed the pre-determined threshold value,the value of the corresponding bit is determined to be unreliable.Thereafter, if even a single bit among the 8 bits, which are decided bythe codes of the soft decision value and group to configure 1 data byte,is determined to be unreliable, the corresponding data byte is marked onthe reliability map as an unreliable data byte.

Herein, determining the reliability of 1 data byte is only exemplary.More specifically, when a plurality of data bytes (e.g., at least 4 databytes) are determined to be unreliable, the corresponding data bytes mayalso be marked as unreliable data bytes within the reliability map.Conversely, when all of the data bits within the 1 data byte aredetermined to be reliable (i.e., when the absolute value of the softdecision values of all 8 bits included in the 1 data byte exceed thepredetermined threshold value), the corresponding data byte is marked tobe a reliable data byte on the reliability map. Similarly, when aplurality of data bytes (e.g., at least 4 data bytes) are determined tobe reliable, the corresponding data bytes may also be marked as reliabledata bytes within the reliability map. The numbers proposed in theabove-described example are merely exemplary and, therefore, do notlimit the scope or spirit of the present invention.

The process of configuring the RS frame and the process of configuringthe reliability map both using the soft decision value may be performedat the same time. Herein, the reliability information within thereliability map is in a one-to-one correspondence with each byte withinthe RS frame. For example, if a RS frame has the size of 540*235 bytes,the reliability map is also configured to have the size of 540*235bytes. FIG. 46( a′) illustrates the process steps of configuring thereliability map according to the present invention. Meanwhile, if a RSframe is configured to have the size of (N+2)235 bytes, the RS framedecoder 907 performs a CRC syndrome checking process on thecorresponding RS frame, thereby verifying whether any error has occurredin each row. Subsequently, as shown in FIG. 46( b), a 2-byte checksum isremoved to configure an RS frame having the size of N*235 bytes. Herein,the presence (or existence) of an error is indicated on an error flagcorresponding to each row. Similarly, since the portion of thereliability map corresponding to the CRC checksum has hardly anyapplicability, this portion is removed so that only N*235 number of thereliability information bytes remain, as shown in FIG. 46( b′).

After performing the CRC syndrome checking process, the RS frame decoder907 performs RS decoding in a column direction. Herein, a RS erasurecorrection process may be performed in accordance with the number of CRCerror flags. More specifically, as shown in FIG. 46( c), the CRC errorflag corresponding to each row within the RS frame is verified.Thereafter, the RS frame decoder 907 determines whether the number ofrows having a CRC error occurring therein is equal to or smaller thanthe maximum number of errors on which the RS erasure correction may beperformed, when performing the RS decoding process in a columndirection. The maximum number of errors corresponds to a number ofparity bytes inserted when performing the RS encoding process. In theembodiment of the present invention, it is assumed that 48 parity byteshave been added to each column.

If the number of rows having the CRC errors occurring therein is smallerthan or equal to the maximum number of errors (i.e., 48 errors accordingto this embodiment) that can be corrected by the RS erasure decodingprocess, a (235,187)−RS erasure decoding process is performed in acolumn direction on the RS frame having 235 N-byte rows, as shown inFIG. 46( d). Thereafter, as shown in FIG. 46( f), the 48-byte paritydata that have been added at the end of each column are removed.Conversely, however, if the number of rows having the CRC errorsoccurring therein is greater than the maximum number of errors (i.e., 48errors) that can be corrected by the RS erasure decoding process, the RSerasure decoding process cannot be performed. In this case, the errormay be corrected by performing a general RS decoding process. Inaddition, the reliability map, which has been created based upon thesoft decision value along with the RS frame, may be used to furtherenhance the error correction ability (or performance) of the presentinvention.

More specifically, the RS frame decoder 907 compares the absolute valueof the soft decision value of the block decoder 905 with thepre-determined threshold value, so as to determine the reliability ofthe bit value decided by the code of the corresponding soft decisionvalue. Also, 8 bits, each being determined by the code of the softdecision value, are grouped to form 1 data byte. Accordingly, thereliability information on this 1 data byte is indicated on thereliability map. Therefore, as shown in FIG. 46( e), even though aparticular row is determined to have an error occurring therein basedupon a CRC syndrome checking process on the particular row, the presentinvention does not assume that all bytes included in the row have errorsoccurring therein. The present invention refers to the reliabilityinformation of the reliability map and sets only the bytes that havebeen determined to be unreliable as erroneous bytes. In other words,with disregard to whether or not a CRC error exists within thecorresponding row, only the bytes that are determined to be unreliablebased upon the reliability map are set as erasure points.

According to another method, when it is determined that CRC errors areincluded in the corresponding row, based upon the result of the CRCsyndrome checking result, only the bytes that are determined by thereliability map to be unreliable are set as errors. More specifically,only the bytes corresponding to the row that is determined to haveerrors included therein and being determined to be unreliable based uponthe reliability information, are set as the erasure points. Thereafter,if the number of error points for each column is smaller than or equalto the maximum number of errors (i.e., 48 errors) that can be correctedby the RS erasure decoding process, an RS erasure decoding process isperformed on the corresponding column. Conversely, if the number oferror points for each column is greater than the maximum number oferrors (i.e., 48 errors) that can be corrected by the RS erasuredecoding process, a general decoding process is performed on thecorresponding column.

More specifically, if the number of rows having CRC errors includedtherein is greater than the maximum number of errors (i.e., 48 errors)that can be corrected by the RS erasure decoding process, either an RSerasure decoding process or a general RS decoding process is performedon a column that is decided based upon the reliability information ofthe reliability map, in accordance with the number of erasure pointswithin the corresponding column. For example, it is assumed that thenumber of rows having CRC errors included therein within the RS frame isgreater than 48. And, it is also assumed that the number of erasurepoints decided based upon the reliability information of the reliabilitymap is indicated as 40 erasure points in the first column and as 50erasure points in the second column. In this case, a (235,187)−RSerasure decoding process is performed on the first column.Alternatively, a (235,187)−RS decoding process is performed on thesecond column. When error correction decoding is performed on all columndirections within the RS frame by using the above-described process, the48-byte parity data which were added at the end of each column areremoved, as shown in FIG. 46( f).

As described above, even though the total number of CRC errorscorresponding to each row within the RS frame is greater than themaximum number of errors that can be corrected by the RS erasuredecoding process, when the number of bytes determined to have a lowreliability level, based upon the reliability information on thereliability map within a particular column, while performing errorcorrection decoding on the particular column. Herein, the differencebetween the general RS decoding process and the RS erasure decodingprocess is the number of errors that can be corrected. Morespecifically, when performing the general RS decoding process, thenumber of errors corresponding to half of the number of parity bytes(i.e., (number of parity bytes)/2) that are inserted during the RSencoding process may be error corrected (e.g., 24 errors may becorrected). Alternatively, when performing the RS erasure decodingprocess, the number of errors corresponding to the number of paritybytes that are inserted during the RS encoding process may be errorcorrected (e.g., 48 errors may be corrected).

After performing the error correction decoding process, as describedabove, a RS frame configured of 187 N-byte rows (or packets) maybeobtained, as shown in FIG. 46( f). Furthermore, the RS frame having thesize of N*187 bytes is sequentially outputted in N number of 187-byteunits. Herein, as shown in FIG. 46( g), the 1-byte MPEG synchronizationbyte that was removed by the transmitting system is added at the end ofeach 187-byte packet, thereby outputting 188-byte mobile service datapackets.

As described above, the digital broadcasting system and the dataprocessing method according to the present invention have the followingadvantages. More specifically, the digital broadcasting receiving systemand method according to the present invention is highly protectedagainst (or resistant to) any error that may occur when transmittingmobile service data through a channel. And, the present invention isalso highly compatible to the conventional receiving system. Moreover,the present invention may also receive the mobile service data withoutany error even in channels having severe ghost effect and noise.Additionally, by inserting known data in a particular position (orplace) within a data region and transmitting the processed data, thereceiving performance of the receiving system may be enhanced even in achannel environment that is liable to frequent changes. Also, bymultiplexing mobile service data with main service data into a burststructure, the power consumption of the receiving system may be reduced.

Moreover, by having the receiving system use the known data informationto perform channel equalization, the channel equalization process may beperformed with more stability. Particularly, by performing differentchannel equalization processes in accordance with the characteristic ofeach region within the data group being divided into a plurality ofregions, the channel equalization process may be performed with morestability in each region, thereby enhancing the receiving performance ofthe present invention. Further, by estimating a remaining carrier phaseerror and by compensating the estimated remaining carrier phase errorfrom the channel-equalized signal, the receiving performance of thepresent invention may be even more enhanced. Finally, the presentinvention is even more effective when applied to mobile and portablereceivers, which are also liable to a frequent change in channel andwhich require protection (or resistance) against intense noise.

It will be apparent to those skilled in the art that variousmodifications and variations can be made in the present inventionwithout departing from the spirit or scope of the inventions. Thus, itis intended that the present invention covers the modifications andvariations of this invention provided they come within the scope of theappended claims and their equivalents.

What is claimed is:
 1. A digital broadcast transmitter comprising: afirst encoder configured to first encode broadcast service data; asecond encoder configured to second encode transmission parameter data;an interleaver configured to interleave the encoded broadcast servicedata and the encoded transmission parameter data; a third encoderconfigured to third encode the interleaved data; and a transmitting unitconfigured to transmit the third encoded data, wherein the third encodeddata are transmitted through data groups and wherein the transmissionparameter data include identification information for identifying acollection of the data groups.
 2. The digital broadcast transmitter ofclaim 1, wherein the transmission parameter data further includeencoding information.
 3. The digital broadcast transmitter of claim 1,wherein the interleaver outputs the data groups including theinterleaved data, each data group having first segments and secondsegments, the first segments having first known data sequences, thesecond segments having second known data sequences, a number of databytes for one of the first known data sequences being different from anumber of data bytes for one of the second known data sequences, Msecond segments being positioned between an Nth first segment and an(N+k)th first segment, L second segments being positioned between the(N+k)th first segment and an (N+k+i)th first segment, and wherein M, L,N, k and i are natural numbers, M≠L.
 4. The digital broadcasttransmitter of claim 3, wherein all of the first known data sequenceshave the same number of data bytes.
 5. The digital broadcast transmitterof claim 3, wherein at least two of the second known data sequences havea different number of data bytes.
 6. The digital broadcast transmitterof claim 3, wherein at least two of the first known data sequences arespaced 16 segments apart.
 7. The digital broadcast transmitter of claim3, wherein at least two of the second known data sequences are spaced 16segments apart.
 8. A method of processing broadcast data in a digitalbroadcast transmitter, the method comprising: first encoding broadcastservice data; second encoding transmission parameter data; interleavingthe encoded broadcast service data and the encoded transmissionparameter data; third encoding the interleaved data; and transmittingthe third encoded data, wherein the third encoded data are transmittedthrough data groups and wherein the transmission parameter data includeidentification information for identifying a collection of the datagroups.
 9. The method of claim 8, wherein the transmission parameterdata further includes encoding information.
 10. The method of claim 8,wherein interleaving the encoded data outputs the data groups includingthe interleaved data, each data group having first segments and secondsegments, the first segments having first known data sequences, thesecond segments having second known data sequences, a number of databytes for one of the first known data sequences being different from anumber of data bytes for one of the second known data sequences, Msecond segments being positioned between an Nth first segment and an(N+k)th first segment, L second segments being positioned between the(N+k)th first segment and an (N+k+i)th first segment, and wherein M, L,N, k and i are natural numbers, M≠L.
 11. The method of claim 10, whereinall of the first known data sequences have the same number of databytes.
 12. The method of claim 10, wherein at least two of the secondknown data sequences have a different number of data bytes.
 13. Themethod of claim 10, wherein at least two of the first known datasequences are spaced 16 segments apart.
 14. The method of claim 10wherein at least two of the second known data sequences are spaced 16segments apart.